Non-isolated DC-DC converters with direct primary to load current

ABSTRACT

DC-DC converters have high side and rectifier circuits, and output capacitor. High side circuit connects between input voltage and output voltage, and has primary winding and auxiliary section that operate transformer properly. Auxiliary may have switches or combination of switches and capacitors. High side circuit converts electrical into magnetic energy through transformer primary, which is then transferred to output through rectifier circuit. It also transfers energy directly to output voltage. Converters have high efficiency, fast dynamic response and high current output. Converters can have large duty cycle and large input voltage and output voltage conversion ratio. High side circuit can be half-bridge, full-bridge or forward converter. Rectifier uses inductors on either side of the secondary, and diodes or synchronous rectifiers, to rectify output voltage. Multi-phase interleaved circuits utilize shared switches to reduce size. High side circuit can utilize resonant tank to decrease switching losses in auxiliary.

CROSS REFERENCE TO RELATED APPLICATIONS

[0001] This application claims priority from and is entitled to thebenefit of the filing date of U.S. Provisional Patent Application No.60/431,740 filed 9 Dec. 2002 under the same title and by the sameinventors.

FIELD OF THE INVENTION

[0002] The invention relates to DC-DC converters. More particularly, itrelates to such converters for low voltage central processing unitapplications.

BACKGROUND OF THE INVENTION

[0003] Computers are widely used both in our personal life and in ourwork. The heart of the computer is the central processing unit (CPU),which performs all the numerical calculation needed for today's everdemanding operation. In order to increase the calculation speed of theCPU, and, thus the speed of the computer, the voltage required to powerthe CPU is becoming lower and lower. At the same time, the current thatthe CPU draws is becoming higher and higher. For example, the nextgeneration CPU will require supply voltage of less than 1V and currentof more than 100A. The current view is that the required CPU supplyvoltage is from 0.8V to 1.6V for the next generation CPU.

[0004] For the CPU in a high-end server, the type of computer used todirect our Internet traffic and data transfer, the CPU's currentrequirement is even larger. The current for a server CPU could be up to500A.

[0005] As an example, in North America, power for a personal computer istypically drawn from a 120V AC wall outlet. This AC voltage is convertedinto a 12V DC voltage by an AC-to-DC converter. The 12V DC isdistributed to a motherboard where the CPU is located. This 12V DCvoltage cannot power the CPU directly. A DC-DC converter (often referredto as a voltage regulator module or VRM) is used to convert the 12V DCinto the lower voltage required by the CPU. This power systemarchitecture is currently preferred from a performance and cost point ofview.

[0006] Another requirement of CPU powering is fast dynamic response.During a time when little calculation is required, the CPU will drawvery low current. For the time when a lot of calculation is required,the CPU will draw large current. The transition between the low currentand large current is very fast. The current change rate can be as highas 10,000A per microsecond. Therefore, the converter should have veryfast dynamic response to meet this requirement.

[0007] If the response speed of the converter is not fast enough, thevoltage across the CPU will have significant overshoot during thetransition from large CPU current to low CPU current because an inductoris typically used in the converter. This voltage overshoot could causedamage to the CPU. Similarly, the voltage across the CPU will havesignificant undershoot during the transition from low CPU current tolarge CPU current. If the voltage is too low, the CPU may not operateproperly.

[0008] In addition, the power loss for the converter should also besmall in order to reduce the temperature rise of the semiconductors usedto implement the converter.

[0009] Referring to FIG. 1, a Buck converter is typically used toconvert incoming 12V into low output voltage such as 1.5V.

[0010] The output voltage of a Buck converter is calculated as:

Vout=D*Vin

[0011] Where Vout is the output voltage and Vin is the input voltage. Dis the duty cycle and is defined as:

D=Ton/Ts

[0012] Where Ton is the time period during which the top switch Q1 isconducting, and Ts is the switching period of Q1.

[0013] In order to achieve Vin=12V to Vout=0.8V conversion, the requiredduty cycle for Buck converter is D=0.8/12=7%. It is noted that a smallduty cycle such as 7% is not optimal for the design and operation of aBuck converter when the switching time of the MOSFET (“metal-oxidesemiconductor field-effect transistor”) is considered. For example, fora typical MOSFET, the turn on time is around 50 ns and the turn off timeis around 100 ns. This means that the MOSFET will be conducting for atleast 150 ns regardless of the control signal. If we assume theswitching frequency is 300 KHz, the switching period is 3.33 μs. Theswitching time of 150 ns is equivalent to 150 ns/3.33 μs=4.5%. Thismeans that we only have control of about 2.5% (7%−4.5%) of conductingtime of the MOSFET. Considering the delay time of the controller, it isvery difficult to design an actual implementation. The compromise is toreduce the switching frequency to a lower level, such as 200 KHz.However, lower switching frequency will also lower the dynamic response,which is a very important performance measurement for DC-DC converters.

[0014] In addition, operating at a very small duty cycle has anotherdetrimental impact to the dynamic response. During the transition fromlow CPU current to high CPU current, the inductor current should beramped up. This can be done by increasing the duty cycle from 7% (takethe above example) to 100% (maximum). The duty cycle has 93% change,which is very beneficial to handle this transition. However, during thetransition from high CPU current to low CPU current, the inductorcurrent should be ramped down. The only way to achieve this is to reducethe duty cycle. Nevertheless, the duty cycle can only be reduced from 7%to 0%. The duty cycle has only 7% change, which results in poor dynamicresponse.

[0015] To improve the dynamic response, it is desirable to select higherswitching frequency for the converter. It is also desirable for theconverter to operate at around 50% duty cycle.

[0016] Referring to FIG. 2, in order to improve the dynamic response aninterleaved Buck converter can be used, such as a four-phase interleavedBuck converter. By interleaving, the equivalent ripple frequency is 4times the switching frequency of each phase. For example, if theswitching frequency of each Buck converter is 200 KHz, the equivalentswitching frequency for four-phase interleaved Buck converter will be800 KHz. Another benefit of interleaving is that the ripple currentthrough the output capacitor and input capacitor is also significantlyreduced. However, each Buck converter still operates at very small dutycycle, which is not desirable.

SUMMARY OF THE INVENTION

[0017] In a first aspect the invention provides a DC-DC converter foruse with a DC power source having a DC voltage across a first voltagesource output and a second voltage source output and with a load. Theconverter includes an input for accepting the DC voltage, the inputhaving a first voltage input and a second voltage input. It alsoincludes an output for outputting a converter DC voltage, the outputhaving a first voltage output and a second voltage output. Furthermoreit includes a high side circuit including a first primary winding of afirst transformer and an auxiliary section, the high side circuitconnected between the first voltage input and the second voltage output.It also includes a rectifier circuit having a first secondary winding ofthe first transformer, the rectifier circuit connected between the firstvoltage output and the second voltage output. There is also an outputcapacitor connected between the first voltage output and the secondvoltage output and across the rectifier circuit.

[0018] In this first aspect an output converter DC voltage between thefirst voltage output and the second voltage output has the same polarityas a DC voltage input between the first voltage input and the secondvoltage input, The auxiliary section is for causing the firsttransformer to transfer power from the first primary winding to thefirst secondary winding and to operate without saturation. The high sidecircuit has a high side circuit output connected such that currentflowing through the first primary winding is directed between the highside circuit output and the first voltage output. The rectifier circuitis for converting output of the first secondary winding into aone-direction waveform and converting the one-direction waveform into aDC voltage. The output capacitor is for filtering the converted DCvoltage.

[0019] The auxiliary section may include switches for repeatedlyconnecting and disconnecting the primary winding from the input, andallowing for resetting of the first transformer. The auxiliary sectionmay have a combination of switches and capacitors. Alternatively, theauxiliary section may have four switches. Each switch may be a MOSFET.

[0020] Alternatively, the auxiliary section may have a first switchconnected between a first side of the first primary winding and thefirst voltage input, a second switch connected between a second side ofthe first primary winding and the first voltage input, a third switchconnected between the first side of the first primary winding and thehigh side circuit output, and a fourth switch connected between thesecond side of the first primary winding and the high side circuitoutput. Each switch may have an input for a gate drive signal forcontrolling the operation of the switch. The gate drive signals mayrepeatedly turn on and turn off the first and fourth switch as well asturn on and turn off the second and third switch.

[0021] Alternatively, the auxiliary section may have a first switchconnected between a first side of the first primary winding and thefirst voltage input, a first capacitor connected between a second sideof the first primary winding and the first voltage input, a secondswitch connected between the first side of the first primary winding andthe high side circuit output, and a second capacitor connected betweenthe second side of the first primary winding and the high side circuitoutput. Each switch may have an input for a gate drive signal forcontrolling the operation of the switch. The converter may include gatedrive signals adapted to repeatedly turn on and turn off the firstswitch and second switch, whereby the first transformer can be resetfrom the capacitors. The capacitors may be large enough that the voltageacross the capacitors will not change significantly during normaloperation of the converter.

[0022] Alternatively, the auxiliary section may have a first switchconnected between a first side of the first primary winding and thefirst voltage input, a first diode connected between a second side ofthe first primary winding and the first voltage input for forwardconduction from the second side of the first primary winding to thefirst voltage input, a second switch connected between the second sideof the first primary winding and the high side circuit output, and asecond diode connected between the first side of the first primarywinding and the high side circuit output for forward conduction from thehigh side circuit output to the first side of the first primary winding.Each switch may have an input for a gate drive signal for controllingthe operation of the switch. The converter may have gate drive signalsadapted to repeatedly turn on and turn off the first switch and secondswitch, whereby the first transformer can be reset by current flowingthrough the first and second diodes.

[0023] Alternatively, the auxiliary section may have a first side of thefirst primary winding connected to the first voltage input, the firstswitch connected between the first side of the first primary winding andthe first side of the first capacitor, the second side of the firstcapacitor connected between the second side of the first switch and thesecond side of the first primary winding, a second switch connectedbetween the second side of the first primary winding and the high sidecircuit output. Each switch may have an input for a gate drive signalfor controlling the operation of the switch. The converter may have gatedrive signals adapted to repeatedly turn on the first switch, whileturning off the second switch, and turn off the first switch, whileturning on the second switch, whereby the first transformer can be resetfrom the first capacitor.

[0024] The rectifier circuit may have a combination of inductors andswitches, wherein the switches are for converting alternating voltage inthe first secondary winding into pulsating one-direction voltage and theinductors are for converting pulsating one-direction voltage into DCvoltage. Alternatively, the rectifier circuit may have a combination ofinductors and diodes, wherein the diodes are for converting pulsatingalternating voltage in the first secondary winding into pulsatingone-direction voltage and the inductors are for converting pulsatingone-direction voltage into DC voltage.

[0025] Alternatively, the rectifier circuit may have a first rectifierswitch connected between the second voltage output and a first side ofthe first secondary winding, a second rectifier switch connected betweena second side of the first secondary winding and the second voltageoutput, a first inductor connected between the first side of the firstsecondary winding and the first voltage output, and a second inductorconnected between the second side of the first secondary winding and thefirst voltage output. Each switch may have an input for a gate drivesignal for controlling the operation of the switch. The converter mayhave gate drive signals adapted to switch the first and second rectifierswitches to convert bi-directional AC voltage at the first secondarywinding into one-direction pulsating voltage.

[0026] Alternatively, the rectifier circuit further may have a firstrectifier switch connected between the second voltage output and a firstside of the first secondary winding, a second rectifier switch connectedbetween a second side of the first secondary winding and the secondvoltage output, and a first inductor connected between the first side ofthe first secondary winding and the first voltage output not in serieswith the second rectifier switch.

[0027] Alternatively, the rectifier circuit may have first and secondrectifier diodes and a first inductor, and the first diode is connectedbetween a first side of the first secondary winding and the firstinductor, and the inductor is further connected between the first diodeand the first voltage output, for forward conduction from the secondarywinding through the inductor, and the second diode is connected between(a) a point between the second side of the first secondary winding andthe second voltage output and (b) a point between the first inductor andfirst diode, also for forward conduction from the secondary windingthrough the inductor. The first and second diodes, the first secondaryand second secondary windings and the inductor may be within a firstrectifier section, and the rectifier circuit also includes a secondrectifier section similar to the first rectifier section, and the firstand second rectifier sections are connected in parallel with one anotherand with the output capacitor and the output

[0028] Alternatively, the rectifier circuit may have a second secondarywinding, first and second rectifier diodes and a first inductor, and asecond side of the first secondary winding is connected to a first sideof the second secondary winding and the second voltage output, and thefirst diode is connected between a first side of the first secondarywinding and the first inductor, and the inductor is further connectedbetween the first diode and the first voltage output, for forwardconduction from the secondary winding through the inductor, and thesecond diode is connected between (a) a point between a second side ofthe second secondary winding and (b) a point between the first inductorand first diode, also for forward conduction from the secondary windingthrough the inductor. The first and second rectifier switches, the firstsecondary and second secondary windings and the inductor may be within afirst rectifier section, and the rectifier circuit further comprises asecond rectifier section similar to the first rectifier section, and thefirst and second rectifier sections are connected in parallel with oneanother and with the output capacitor and the output.

[0029] Alternatively, the rectifier circuit may have a second secondarywinding, first and second rectifier switches and a first inductor, and asecond side of the first secondary winding is connected to a first sideof the second secondary winding and the inductor which is furtherconnected to the first voltage output, and the first rectifier switch isconnected between a first side of the first secondary winding and thesecond voltage output, and the second rectifier switch is connectedbetween a second side of the second secondary winding and the secondvoltage output. The first and second rectifier switches, the firstsecondary winding and the first and second inductors may be within afirst rectifier section, and the rectifier circuit also includes asecond rectifier section similar to the first rectifier section, and thefirst and second rectifier sections are connected in parallel with oneanother and with the output capacitor and the output.

[0030] The converter may have a second converter similar to the firstconverter, wherein the two converters are connected in parallel with oneanother at their respective inputs and outputs. The output capacitors ofthe two converters may be combined as a single physical capacitor. Thetwo converters may have interleaved gate drive signals, whereby currentripple incoming to the output capacitor is reduced, allowing forreduction in the size of the output capacitor.

[0031] The converter may have a second transformer and the high sidecircuit may have a second primary winding of the second transformer, andfirst and second second primary switches, wherein the first secondprimary switch is connected between the first voltage input and a firstside of the second primary winding, and the second second primary switchis connected between the first side of the second primary winding andthe high side circuit output, and a second side of the second primarywinding is connected to a side of the first primary winding, and therectifier circuit may have a second rectifier circuit similar to andconnected in parallel with the first rectifier circuit, wherein thesecond rectifier circuit includes a second second rectifier secondarywinding of the second transformer. The second primary switches may haveinput for gate drive signals for operating the second primary windingout of phase with the first primary winding.

[0032] The converter may have a second transformer and the high sidecircuit may have a second primary winding of the second transformer, andfirst and second second primary switches, wherein the first secondprimary switch is connected between the first voltage input and a firstside of the second primary winding, and the second second primary switchis connected between the first side of the second primary winding andthe high side circuit output, and a second side of the second primarywinding is connected to a side of the first primary winding, and therectifier circuit may have a second rectifier secondary winding, asecond rectifier inductor and a second rectifier switch, wherein a firstside of the second rectifier secondary winding is connected to a firstside of the first secondary winding and the second rectifier switch isconnected between a second side of the second secondary winding and thesecond voltage output, and the inductor is connected between the secondside of the second secondary winding and the high side circuit output,not in series with the second rectifier switch. The second primaryswitches may have inputs for gate drive signals for operating the secondprimary winding out of phase with the first primary winding, and thefirst and second rectifier circuit have inputs for gate drive signalsfor operating the rectifier circuit secondary windings phase shiftedfrom one another.

[0033] The converter may have a second transformer and a thirdtransformer, and the high side circuit may have a second primary windingof the second transformer, and first and second second primary switches,wherein the first second primary switch is connected between the firstvoltage input and a first side of the second primary winding, and thesecond second primary switch is connected between the first side of thesecond primary winding and the high side circuit output, and a secondside of the second primary winding is connected to a side of the firstprimary winding, and the rectifier circuit may have a second rectifiercircuit and a third rectifier circuit each similar to and connected inparallel with the first rectifier circuit, wherein the second rectifiercircuit includes a second second rectifier secondary winding of thesecond transformer and the third rectifier circuit includes a thirdthird rectifier secondary winding of the third transformer.

[0034] The second primary switches may have input for gate drive signalsfor operating the second primary winding out of phase with the firstprimary winding, and no additional drive components are added for thethird primary winding, wherein the converter has gate drive inputs foroperating the third primary winding partially in phase with the firstprimary winding and partially in phase with the second primary winding.

[0035] The first voltage input may be for accepting a DC potential thatis positive when compared to a DC potential for acceptance by the secondvoltage input. The DC potential of the first voltage output may bepositive when compared to the DC potential of the second voltage output.

[0036] The input voltage of the converter may be within a range of 10.8volts DC to 13.2 volts DC, and the out put voltage is within a range of0.8 volts DC to 1.6 volts DC.

[0037] The duty cycle of single phase and two-phase converters may bebetween 40% and 60%. The duty cycle may be approximately 50%. The dutycycle of three-phase converters may be approximately 33⅓%.

[0038] The converter may have a second high side circuit similar to thefirst high side circuit, connected in parallel with the first high sidecircuit, and a second rectifier circuit connected in parallel with thefirst rectifier circuit. The converter may have inputs for drive signalsto operate the first high side circuit and the first rectifier circuitout of phase with the second high side circuit and the second rectifiercircuit, respectively. The converter may have gate drive signals foroperating the first high side circuit and the first rectifier circuitout of phase with the second high side circuit and the second rectifiercircuit, respectively.

[0039] The converter may have a current sensor in series with the highside circuit. Current sensed at the current sensor may be for use indetermining the timing of gate drive signals for operating the high sidecircuit. The current sensed at the current sensor may also be used tomake current sharing between two or more parallel connected converters.

[0040] The output of the first secondary winding may be a pulsatingvoltage and the one-direction waveform may be a one-direction voltage.

[0041] The high side circuit may have a resonant tank.

[0042] The resonant tank may have a first capacitor in parallel with thefirst primary winding and a first inductor in series with the firstprimary winding between the first primary winding and the auxiliarysection.

[0043] The resonant tank further may also have a second capacitor inseries with the first inductor between the first primary winding and theauxiliary section.

[0044] The resonant tank may also have a second inductor in parallelwith the first primary winding and the first capacitor.

[0045] The resonant tank may have a first inductor and a first capacitorin series with one another between the first primary winding and theauxiliary section. In this case, the rectifier circuit may have a dutycycle and the duty cycle may be altered to change the output voltage ofthe converter.

[0046] For a resonant tank converter, the rectifier circuit may have afull-bridge rectifier. In such case, switches within the rectifiercircuit may be controlled by phase control to regulate output voltage.

[0047] The rectifier circuit may have a half-bridge rectifier.

[0048] Switches in the resonant tank converter may be controlled byswitching frequency control to regulate output voltage.

[0049] The auxiliary section may be a full-bridge auxiliary section. Insuch case for a resonant tank converter, switches within the auxiliarysection may be controlled by phase shift control to regulate outputvoltage.

[0050] In another aspect the invention provides a method of operating aconverter of the type described above including the steps of driving theauxiliary section to cause the first transformer to transfer power fromthe first primary winding to the first secondary winding, while at thesame time driving the auxiliary section to cause the transformer tooperate without saturation.

[0051] Other aspects of the invention, including other methods employingconverters, will be evident from the figures and detailed descriptionherein.

BRIEF DESCRIPTION OF THE DRAWINGS

[0052] For a better understanding of the present invention and to showmore clearly how it may be carried into effect, reference will now bemade, by way of example, to the accompanying drawings which show thepreferred embodiment of the present invention and in which:

[0053]FIG. 1 is a schematic diagram of a Buck converter of knownconfiguration;

[0054]FIG. 2 is a schematic diagram of a four-phase interleaved Buckconverter of known configuration;

[0055]FIG. 3 is a schematic diagram of a non-isolated DC converter inaccordance with a preferred embodiment of the present invention;

[0056]FIG. 4 is a schematic diagram of one embodiment of a high sidecircuit for the converter of FIG. 3;

[0057]FIG. 5 is a schematic diagram of gate drive signals for Q1 and Q2for the embodiment shown in FIG. 4;

[0058]FIG. 6 is a schematic diagram of another embodiment of a high sidecircuit for the converter of FIG. 3;

[0059]FIG. 7 is a schematic diagram of gate drive signals for Q1 and Q2for the embodiment shown in FIG. 6;

[0060]FIG. 8 is a schematic diagram of another embodiment of a high sidecircuit for the converter of FIG. 3 using two MOSFETs and two diodes;

[0061]FIG. 9 is a schematic diagram of another embodiment of a high sidecircuit for the converter of FIG. 3;

[0062]FIG. 10 is a schematic diagram of an embodiment of a rectifiercircuit for the converter of FIG. 3 using one transformer winding andsynchronous rectifiers;

[0063]FIG. 11 is a schematic diagram of another embodiment of arectifier circuit for the converter of FIG. 3 using one transformerwinding and synchronous rectifiers;

[0064]FIG. 12 is a schematic diagram of another embodiment of arectifier circuit for the converter of FIG. 3 using one transformerwinding and diodes;

[0065]FIG. 13 is a schematic diagram of another embodiment of arectifier circuit for the converter of FIG. 3 using two windings and twodiodes;

[0066]FIG. 14 is a schematic diagram of another embodiment of arectifier circuit for the converter of FIG. 3 using two windings andsynchronous rectifiers;

[0067]FIG. 15 is a schematic diagram of a non-isolated full-bridge DCconverter embodiment of the converter of FIG. 3;

[0068]FIG. 16 is a schematic diagram of an equivalent circuit for theconverter of FIG. 15 when Q1, Q4 and Q6 are on;

[0069]FIG. 17 is a schematic diagram of an equivalent circuit for theconverter of FIG. 15 when Q1, Q2, Q3 and Q4 are off and Q5, Q6 are on;

[0070]FIG. 18 is a schematic diagram of an equivalent circuit for theconverter of FIG. 15 when Q2, Q3 and Q5 are on;

[0071]FIG. 19 is a graphic illustration of certain waveforms used withthe non-isolated full-bridge DC converter of FIG. 15 for regular PWM(pulse width modulated) gate drive;

[0072]FIG. 20 is a graphic illustration of certain waveforms used withthe non-isolated full-bridge DC converter of FIG. 15 for phase-shiftedPWM gate drive signals for Q1, Q2, Q3, Q4, Q5 and Q6;

[0073]FIG. 21 is a schematic diagram of another non-isolated full-bridgeDC converter embodiment of the converter of FIG. 15;

[0074]FIG. 22 is a schematic diagram illustrating the use of synchronousrectifiers instead of diodes in the converter of FIG. 21;

[0075]FIG. 23 is a schematic diagram of a non-isolated full-bridge DCconverter embodiment of the converter of FIG. 3 with two rectifiersections;

[0076]FIG. 24 is a schematic diagram of another non-isolated full-bridgeDC converter embodiment of the converter of FIG. 3 with two rectifiersections;

[0077]FIG. 25A is a schematic diagram illustrating the use ofsynchronous rectifiers instead of diodes in the embodiment of FIG. 24;

[0078]FIG. 25B is a schematic diagram of a two-phase interleavednon-isolated DC converter embodiment of the converter of FIG. 3;

[0079]FIG. 26 is a schematic diagram of a two-phase interleavednon-isolated DC converter embodiment of the converter of FIG. 3;

[0080]FIG. 27 is a graphic illustration of certain waveforms used withthe two-phase interleaved non-isolated full-bridge DC converterembodiment of FIG. 26 with duty cycle of 40%;

[0081]FIG. 28 is a graphic illustration of certain waveforms used withthe two-phase interleaved non-isolated full-bridge DC converterembodiment of FIG. 26 with duty cycle of 50%;

[0082]FIG. 29 is a graphic illustration of certain waveforms used withthe two-phase interleaved non-isolated full-bridge DC converterembodiment of FIG. 26 with duty cycle of 60%;

[0083]FIG. 30 is a schematic diagram of a two-phase interleavednon-isolated full-bridge DC converter embodiment of the converter ofFIG. 3 with a high side circuit shared switch;

[0084]FIG. 31 is a schematic diagram of gate drive signals for the DCconverter of FIG. 30;

[0085]FIG. 32 is a graphic illustration of certain waveforms used withthe two-phase interleaved non-isolated full-bridge DC converterembodiment of FIG. 30 with duty cycle of 40%;

[0086]FIG. 33 is a graphic illustration of certain waveforms used withthe two-phase interleaved non-isolated full-bridge DC converterembodiment of FIG. 30 with duty cycle of 50%;

[0087]FIG. 34 is a graphic illustration of certain waveforms used withthe two-phase interleaved non-isolated full-bridge DC converterembodiment of FIG. 30 with duty cycle of 60%;

[0088]FIG. 35A is a schematic diagram of a two-phase interleavednon-isolated full-bridge DC converter embodiment of the converter ofFIG. 3 with high side circuit shared switch and rectifier circuit sharedswitch;

[0089]FIG. 35B is a schematic diagram of a basic three-phase interleavednon-isolated full-bridge DC converter embodiment of the converter ofFIG. 3

[0090]FIG. 36 is a schematic diagram of a three-phase interleavednon-isolated full-bridge DC converter embodiment of the converter ofFIG. 3 with primary shared switches;

[0091]FIG. 37A and FIG. 37B are each a graphic illustration of a gatedrive scheme for the converter of FIG. 36 with a duty cycle of 30%;

[0092]FIG. 38A is a schematic diagram of a simplified three-phaseinterleaved non-isolated full-bridge DC converter embodiment of theconverter of FIG. 3 with primary shared switches;

[0093]FIG. 38B is a graphic illustration of one gate drive scheme forthe converter shown in FIG. 38A.

[0094]FIG. 39 is a schematic diagram of a simplified three-phaseinterleaved non-isolated full-bridge DC converter embodiment of theconverter of FIG. 3 with primary shared switches and rectifier circuitshared switch;

[0095]FIG. 40 is a schematic diagram of a non-isolated half-bridge DCconverter embodiment of the converter of FIG. 3;

[0096]FIG. 41 is a schematic diagram of an equivalent circuit for theconverter of FIG. 40 when Q1, Q4 and Q6 are on;

[0097]FIG. 42 is a schematic diagram of an equivalent circuit for theconverter of FIG. 40 when Q3 and Q4 are on;

[0098]FIG. 43 is a schematic diagram of an equivalent circuit for theconverter of FIG. 40 when Q2 and Q4 are on;

[0099]FIG. 44 is a graphic illustration of certain waveforms for thehalf-bridge converter of FIG. 40;

[0100]FIG. 45 is a schematic diagram of another embodiment of anon-isolated half-bridge DC converter embodiment of the converter ofFIG. 3;

[0101]FIG. 46 is a schematic diagram illustrating the use of asynchronous rectifiers instead of diodes in the converter of FIG. 45;

[0102]FIG. 47 is a schematic diagram of a non-isolated half-bridge DCconverter embodiment of the converter of FIG. 3 with two rectifiersections;

[0103]FIG. 48 is a schematic diagram of a two-phase interleavednon-isolated half-bridge DC converter embodiment of the converter ofFIG. 3;

[0104]FIG. 49 is a graphic illustration of gate drive signals otherwaveforms for the two-phase interleaved non-isolated half-bridge DCconverter embodiment of FIG. 48 with a duty cycle of 40%;

[0105]FIG. 50 is a schematic diagram of a two-phase interleavedhalf-bridge DC converter embodiment of the converter of FIG. 3 withsecondary shared switches;

[0106]FIG. 51 is a schematic diagram of a non-isolated forward DCconverter embodiment of the converter of FIG. 3;

[0107]FIG. 52 is a schematic diagram of the non-isolated forward DCconverter embodiment shown in FIG. 51 when synchronous rectifiers areused;

[0108]FIG. 53 is a schematic diagram of a non-isolated forward DCconverter embodiment of FIG. 3 when the rectifier circuit shown in FIG.10 is used;

[0109]FIG. 54 is a schematic diagram of another non-isolated forward DCconverter embodiment of the converter of FIG. 3;

[0110]FIG. 55 is a schematic diagram of the non-isolated forward DCconverter embodiment shown in FIG. 54 when synchronous rectifiers areused;

[0111]FIG. 56 is a schematic diagram of the non-isolated forward DCconverter embodiment of FIG. 54 when the rectifier circuit shown in FIG.10 is used;

[0112]FIG. 57 is a schematic diagram of another non-isolated forward DCconverter embodiment of the converter of FIG. 3 when a third winding isused to reset the transformer core;

[0113]FIG. 58 is a schematic diagram of a two-phase interleavednon-isolated forward DC converter embodiment of the circuit shown inFIG. 51;

[0114]FIG. 59 is a schematic diagram of a two-phase interleavednon-isolated forward DC converter embodiment of the circuit shown inFIG. 52;

[0115]FIG. 60 is a schematic diagram of a two-phase interleavednon-isolated forward DC converter embodiment of the circuit shown inFIG. 53;

[0116]FIG. 61 is a schematic diagram of a two-phase interleavednon-isolated forward DC converter embodiment of the circuit shown inFIG. 54

[0117]FIG. 62 is a schematic diagram of a two-phase interleavednon-isolated forward DC converter embodiment of the circuit shown inFIG. 55;

[0118]FIG. 63 is a schematic diagram of a two-phase interleavednon-isolated forward DC converter embodiment of the circuit shown inFIG. 56;

[0119]FIG. 64 is a general schematic diagram of two-phase interleavednon-isolated DC converter for current sensing and current sharing;

[0120]FIG. 65 is a schematic diagram showing one implementation ofcurrent sensing circuit for two-phase interleaved full-bridge DCconverter;

[0121]FIG. 66 is a schematic diagram showing one implementation ofcurrent sensing circuit for two-phase interleaved half-bridge DCconverter;

[0122]FIG. 67 is a schematic diagram showing one implementation ofcurrent sensing circuit for two-phase interleaved forward DC converter;

[0123]FIG. 68A is a block diagram of a non-isolated DC converter inaccordance with an alternate preferred embodiment of the presentinvention;

[0124]FIG. 68B is a schematic diagram of a non-isolated parallelresonant full-bridge DC-DC converter embodiment of the converter of FIG.68A with full-bridge rectifier;

[0125]FIG. 69 is a schematic diagram of a non-isolated parallel resonantfull-bridge DC-DC converter embodiment of the converter of FIG. 68A withcurrent doubler rectifier;

[0126]FIG. 70 is a schematic diagram of a non-isolated parallel resonantfull-bridge DC-DC converter embodiment of the converter of FIG. 68A withcenter tapped transformer rectifier;

[0127]FIG. 71 is a schematic diagram of a non-isolated parallel resonantfull-bridge DC-DC converter embodiment of the converter of FIG. 68A withphase control rectifier;

[0128]FIG. 72 is a schematic diagram of a non-isolated parallel resonantfull-bridge DC-DC converter embodiment of the converter of FIG. 68A withLCC resonant tank and full-bridge rectifier;

[0129]FIG. 73 is a schematic diagram of a non-isolated parallel resonantfull-bridge DC-DC converter embodiment of the converter of FIG. 68A withLCC resonant tank and current doubler rectifier;

[0130]FIG. 74 is a schematic diagram of a non-isolated parallel resonantfull-bridge DC-DC converter embodiment of the converter of FIG. 68A withLCC resonant tank and center tapped transformer rectifier;

[0131]FIG. 75 is a schematic diagram of a non-isolated parallel resonantfull-bridge DC-DC converter embodiment of the converter of FIG. 68A withLCC resonant tank and phase control rectifier;

[0132]FIG. 76 is a schematic diagram of a non-isolated parallel resonantfull-bridge DC-DC converter embodiment of the converter of FIG. 68A withLCLC resonant tank and full-bridge rectifier;

[0133]FIG. 77 is a schematic diagram of a non-isolated parallel resonantfull-bridge DC-DC converter embodiment of the converter of FIG. 68A withLCLC resonant tank and current doubler rectifier;

[0134]FIG. 78 is a schematic diagram of a non-isolated parallel resonantfull-bridge DC-DC converter embodiment of the converter of FIG. 68A withLCLC resonant tank and center tapped transformer rectifier;

[0135]FIG. 79 is a schematic diagram of a non-isolated parallel resonantfull-bridge DC-DC converter embodiment of the converter of FIG. 68A withLCLC resonant tank and phase control rectifier;

[0136]FIG. 80 is a schematic diagram of a non-isolated series resonantfull-bridge DC-DC converter embodiment of the converter of FIG. 68A withfull-bridge rectifier;

[0137]FIG. 81 is a schematic diagram of a full-bridge series resonantconverter embodiment of the converter of FIG. 68A with center tappedtransformer rectifier;

[0138]FIG. 82 is a schematic diagram of a non-isolated parallel resonanthalf-bridge DC-DC converter embodiment of the converter of FIG. 68A withfull-bridge rectifier;

[0139]FIG. 83 is a schematic diagram of a non-isolated parallel pesonanthalf-bridge DC-DC Converter embodiment of the converter of FIG. 68A withcurrent doubler rectifier;

[0140]FIG. 84 is a schematic diagram of a non-isolated parallel resonanthalf-bridge DC-DC converter embodiment of the converter of FIG. 68A withcenter tapped transformer rectifier;

[0141]FIG. 85 is a schematic diagram of a non-isolated parallel resonanthalf-bridge DC-DC converter embodiment of the converter of FIG. 68A withphase control rectifier;

[0142]FIG. 86 is a schematic diagram of a non-isolated parallel resonanthalf-bridge DC-D Converter embodiment of the converter of FIG. 68A withLCC resonant tank and full-bridge rectifier;

[0143]FIG. 87 is a schematic diagram of a non-isolated parallel resonanthalf-bridge DC-DC converter embodiment of the converter of FIG. 68A withLCC resonant tank and current doubler rectifier;

[0144]FIG. 88 is a schematic diagram of a non-isolated parallel resonanthalf-bridge DC-DC converter embodiment of the converter of FIG. 68A withLCC resonant tank and center tapped transformer rectifier;

[0145]FIG. 89 is a schematic diagram of a non-isolated parallel resonanthalf-bridge DC-DC converter embodiment of the converter of FIG. 68A withLCC resonant tank and phase control rectifier;

[0146]FIG. 90 is a schematic diagram of a non-isolated parallel resonanthalf-bridge DC-DC converter embodiment of the converter of FIG. 68A withLCLC resonant tank and full-bridge rectifier;

[0147]FIG. 91 is a schematic diagram of a non-isolated parallel resonanthalf-bridge DC-DC converter embodiment of the converter of FIG. 68A withLCLC resonant tank and current doubler rectifier;

[0148]FIG. 92 is a schematic diagram of a non-isolated parallel resonanthalf-bridge DC-DC converter embodiment of the converter of FIG. 68A withLCLC resonant tank and center tapped transformer rectifier;

[0149]FIG. 93 is a schematic diagram of a non-isolated parallel resonanthalf-bridge DC-DC converter embodiment of the converter of FIG. 68A withLCLC resonant tank and phase control rectifier;

[0150]FIG. 94 is a schematic diagram of a non-isolated series resonanthalf-bridge DC-DC converter embodiment of the converter of FIG. 68A withfull-bridge rectifier; and

[0151]FIG. 95 is a schematic diagram of a non-isolated series resonanthalf-bridge DC-DC converter embodiment of the converter of FIG. 68A withcenter tapped transformer rectifier.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT

[0152] Referring to FIG. 3, a non-isolated DC converter 301 (outlined indashed lines) consists of three blocks: high side circuit 305, rectifiercircuit 307 and output filter capacitor Co. The high side circuit 305connects directly between positive point V_(in+) of the input voltageV_(in) and the positive point V_(out+) of the output V_(out). The highside circuit 305 is therefore not isolated from the rectifier circuit307 and, thus, the converter is “non-isolated”.

[0153] As will be discussed, the high side circuit 305 includes one ormore transformer primary winding(s) and an auxiliary section that willoperate the transformer(s) properly. This auxiliary section may consistof switches (such as MOSFET, BJT (“bipolar junction transistor”), etc).This auxiliary section may also consist of a combination of switches andcapacitors. The function of this auxiliary section is to make thetransformer operate properly. The term “operate properly” means that thetransformer will not saturate and the power can be transferred from eachprimary winding to its secondary winding(s) efficiently. Variousexamples of auxiliary sections, such as full-bridge, half-bridge, andforward will be described herein.

[0154] The high side circuit 305 serves two functions. One is that itconverts the electrical energy into magnetic energy through thetransformer primary winding(s). The other function is to transfer energydirectly to the output voltage V_(out).

[0155] The converter 301 can be used to meet the stringent powerrequirement of the next generation CPU. The circuit topologies describedherein are able to achieve high efficiency, fast dynamic response andprovide high current to the CPU.

[0156] Also to be described are methods of operating the non-isolated DCconverter 301. Using these methods, the non-isolated DC converter willoperate with a large duty cycle and at the same time, have a large inputvoltage to output voltage conversion ratio.

[0157] Referring to FIG. 4, one embodiment 401 of the high side circuit305 consists of one primary winding of transformer T1A and four MOSFETsQ1, Q2, Q3, Q4. The left point A of the high side circuit 401 isconnected to Vin and the right point B of the high side circuit 401 isconnected to Vout.

[0158] Gate drive signals of Q1, Q2, Q3, and Q4 should be arranged sothat transformer T1A can operate properly. Referring to FIG. 5, oneexample of such as arrangement is shown as Vgs1, Vgs2, Vgs3 and Vgs4.The gate signals Vgs1 and Vgs4 for Q1 and Q4 are the same and gatesignals Vgs2 and Vgs3 for Q2 and Q3 are the same. In the figure, a highvoltage level means the MOSFET is turned on and a low voltage levelmeans the MOSFET is turned off.

[0159] Referring to FIG. 6, another embodiment 601 of the high sidecircuit 305 is shown. The high side circuit 601 consists of onetransformer primary winding T1A, two MOSFETs Q1, Q2 and two capacitorsC1, C2. The left point A of the high side circuit 601 is connected toVin and the right point B of the high side circuit is connected to Vout.

[0160] The value of C1 and C2 should be large enough so that the voltageacross C1 and C2 does not change significantly during normal operationof the circuit 601. In other words, the voltage across C1 and C2 is a DCvoltage with small voltage ripple. The gate drive of Q1 and Q2 should bearranged so that transformer T1A can operate properly. Referring to FIG.7, one example of such as arrangement is shown as Vgs1 and Vgs2.

[0161] Referring to FIG. 8, another embodiment 801 of the high sidecircuit 305 is shown. This embodiment 801 consists of one transformerprimary winding T1A, two MOSFETs Q1,Q2 and two diodes D1, D2. The leftside A of the high side circuit 801 is connected to the input voltageVin. The right side B of the high side circuit 801 is connected to theoutput voltage Vout.

[0162] Q1 and Q2 are turned on at same time. When Q1 and Q2 are off,magnetizing current flows through D1 and D2 from point B to point A andthus resets the core of transformer T1.

[0163] Referring to FIG. 9, another embodiment 901 of the high sidecircuit 305 is illustrated using one transformer primary winding T1A,two MOSFETs Q1,Q2 and one capacitor C1. Point A is connected to theinput voltage Vin and point B is connected to the output voltage Vout.

[0164] It is noted that MOSFET Q1 carries input current. MOSFET Q2carries only the magnetizing current. When Q1 is on, Q2 is off. When Q1is off, Q2 is on. The voltage developed across C1 is used to reset thecore of transformer T1.

[0165] It is noted that there are other embodiments of the high sidecircuit 305 as will be evident to one skilled in the art using thisdescription. Not all of them have been described, nor will they be. Someadditional examples will be given later.

[0166] Returning to FIG. 3, the rectifier circuit 307 is connected inparallel with the output voltage Vout and in parallel with the outputcapacitor Co. It includes one or more transformer secondary winding(s).It includes diodes that will convert the bi-directional voltage from thetransformer secondary winding into a pulsating, one-direction voltage.In addition, it consists of one or more inductors that will convert thepulsating, one-direction voltage through the secondary winding(s) into aDC voltage.

[0167] It is noted that the diodes can be replaced by MOSFETs in orderto reduce the power loss.

[0168] The function of the rectifier circuit 307 is to convert the ACvoltage from the transformer secondary winding(s) into a DC voltageVout. It also provides a portion of the total load current.

[0169] Referring to FIG. 10, one embodiment 1001 of the rectifiercircuit 307 is shown. In this case, the rectifier circuit 1001 includesone transformer secondary winding T, two MOSFETs Q1,Q2 and two inductorsL1,L2. The top point C is connected to the positive point Vout+ ofoutput voltage Vout. The bottom point D is connected to negative pointVout− of the output voltage Vout. In this embodiment, MOSFETs Q1 and Q2are used as synchronous rectifiers to reduce power loss. The waveform ofthe gate drive signals Vgs1, Vgs2 for Q1 and Q2 will depend on theconnection of the high side circuit 305. The objective of Q1 and Q2 isto convert bi-directional AC voltage that appears at the transformersecondary winding T into one-direction pulsating voltage.

[0170] Referring to FIG. 11, another embodiment 1101 of the rectifiercircuit 307 is shown. In this case, the rectifier circuit 1101 consistsof one transformer secondary winding T1B, two MOSFETs Q1,Q2 and oneinductor L. The top point C is connected to the positive point Vout+ ofoutput voltage Vout. The bottom point D is connected to negative pointVout− of the output voltage Vout. MOSFETs Q1,Q2 are used as synchronousrectifiers to reduce the power loss. The waveform of the gate drivesignal Vgs1, Vgs2 for Q1 and Q2 will depend on the connection of thehigh side circuit 305. The objective of Q1 and Q2 is to convert thebi-directional AC voltage that appears at the transformer secondarywinding T1B into one-direction pulsating voltage.

[0171] Referring to FIG. 12, when the MOSFETs Q1, Q2 in FIG. 11 arereplaced by diodes D1, D2, another embodiment 1201 of the rectifiercircuit 307 is obtained. In this figure, the diodes D1, D2 are used toconvert the bi-directional voltage across the transformer secondarywinding T1B into one-direction pulsating voltage. Inductor L is used tofilter the pulsating voltage into DC voltage Vout

[0172] Some rectifier circuits 307 consist of two transformer secondarywindings. Referring to FIG. 13, a rectifier circuit 1301 with twotransformer secondary windings T1B, T1C is shown. It uses two diodes D1,D2 as rectifier switches. It uses only one inductor L.

[0173] Referring to FIG. 14, another rectifier circuit 1401 with twosecondary windings T1B, T1C and synchronous rectifiers Q1, Q2 is shown.It requires one inductor L.

[0174] As will be evident to those skilled in the art using thisdescription, there are other embodiments of the rectifier circuit 307that have not been described herein.

[0175] Referring again to FIG. 3, the output capacitor Co is connectedin parallel with the load circuit LOAD. It is also connected in parallelwith the rectifier circuit 307. The function of the output capacitor Cois to absorb the pulsating current that comes from the high side circuit305. The value of the output capacitor Co should be large enough toprovide a DC voltage across the load.

[0176] If the capacitor Co is not used, the output voltage Vout may havesignificant ripple voltage such that the load LOAD may not operateproperly.

[0177] It should be noted that the high side circuits described aboveonly show the basic operation and requirement of the high side circuit305 as shown in FIG. 3. In actual implementation, some modification ofthe above high side circuits can be added to further improve theoperation of the high side circuits. For example, leakage inductance ofthe transformer can be intentionally designed larger to make the MOSFETswitches operate at more favourable conditions. A snubber circuit can beadded to reduce the power dissipated in the MOSFETs, diode, ortransformer. Some other auxiliary circuit can be added around theMOSFETs to make the operation of these MOSFETs more favourable, such asreducing the switching loss.

[0178] It should also be noted that the rectifier circuits describedabove only show the basic operation and requirement of the rectifiercircuit 307 as shown in FIG. 3. In actual implementation, somemodification of the above rectifier circuits can be added to furtherimprove operation of the rectifier circuits. For example, the gate drivesignals for the synchronous rectifier switches used in the rectifiercircuit 307 can be derived from the transformer secondary winding tosimplify the gate drive circuit. Other gate drive timing circuits can beadded to further improve the operation of the synchronous rectifier. Asnubber circuit may also be added across the synchronous rectifiers ordiodes to reduce the power dissipation.

[0179] In addition, it should be noted that only one capacitor Co isused as the output filter to smooth output voltage. In actualimplementation, other additional filters can be added after Co. Forexample, a pi filter can be used to further reduce the output voltageripple. Similarly, no input filter is shown in the circuit diagram ofFIG. 3. In actual implementation, at least a capacitor is needed toprovide the input ripple current so that the DC source does not need toprovide the ripple current. This is usually the preferredimplementation. Additional EMI filters can be used to reduce the noiseinjected into input voltage source.

[0180] In converter 301, as shown in FIG. 3, the high side circuit 305is connected between the positive point of the input voltage (Vin+) andthe positive point of the output voltage (Vout+). It should be notedthat the converter 301 will also operate properly when the high sidecircuit is connected between the negative point of the input voltage(Vin−) and the negative point of the output voltage (Vout−). In thisimplementation, different components might be used for the high sidecircuit. The connection of the components might also be changed to makethe high side circuit operate properly. Based on the fact that in actualimplementation, it is preferred to put the high side circuit between thepositive points of Vin and Vout, the following description assumes thatconnection.

[0181] Nevertheless, the circuit 301 will operate when the high sidecircuit 305 is connected between the negative points of Vin and Vout.

[0182] Thus, the voltage inputs to the converters could simply be afirst voltage and a second voltage, rather than always being positiveand negative DC voltages (or “potentials”) with the positive voltagealways being the input voltage to the high side circuit. Similarly, theoutput voltage of the converter could be first and second DC voltages,rather than requiring the converter output that receives current fromthe high side circuit to be a positive voltage when compared to theother converter output. In any case the output converter DC voltagebetween the first voltage and the second voltage has the same polarityas a DC voltage input between the first voltage and the second voltage.

[0183] Accordingly, the term “high side circuit” is used herein althoughthe high side circuit may be in the low side of the converter. The “highside circuit” could alternatively be referred to as a “primary side” or“primary side circuit”. Similarly, “rectifier circuit” couldalternatively be referred to as a “secondary side” or “secondary sidecircuit”.

[0184] All of the above variations and any other variations toconverters employing the principles described herein are included withinthe scope of those principles and the invention as later claimed.

[0185] Several non-isolated full-bridge DC converters will be described.In these DC converters, the high side circuit 301 includes four MOSFETsas switches and one transformer primary winding, as shown in FIG. 3. Anembodiment of a rectifier circuit 307 using multiple rectifier sections(duplicated rectifier circuits) connected in parallel will also bediscussed.

[0186] Referring to FIG. 15, a non-isolated full-bridge DC converter1501 is shown.

[0187] The full-bridge converter 1501 topology includes six MOSFETsQ1-Q6, one transformer T1, with one primary winding T1A and onesecondary side winding T1B, two inductors L1, L2 and one outputcapacitor Co. The operation of the circuit 1501 will be discussed laterbelow.

[0188] Q1, Q2, Q3 and Q4 are primary switches. They can be implemented,for example, by IRL7467 from International Rectifier. This is true forall the primary switches described herein. The drains of Q1 and Q3 areconnected together and then connected to the positive terminal of theinput voltage Vin+. The source of Q1 and the drain of Q2 are connectedtogether. The source of Q3 and the drain of Q4 are connected together.The source of Q2 and Q4 are connected together and then connected to theoutput voltage terminal Vout+.

[0189] Q5 and Q6 are synchronous rectifier switches. They can bereplaced by diodes. Q5 and Q6 can be implemented by, for example,IRLR8103 from International Rectifier. This is true for each of thesecondary MOSFETs described herein. The sources of Q5 and Q6 areconnected together and then are connected to ground point (or negativeterminal of input voltage, which is the same point as negative terminalof output voltage Vout−). The drain of Q5 is connected with one terminalof inductor L1. The other terminal of L1 is connected to the positivepoint of output voltage Vout+. The drain of Q6 is connected with oneterminal of L2. The other terminal of L2 is connected to the positiveterminal of the output voltage Vout+.

[0190] The primary winding T1A is connected between the drain of Q2 anddrain of Q4. The secondary winding T1B of transformer T1 is connectedbetween drain of Q5 and drain of Q6.

[0191] For VRM applications, Q1, Q2, Q3 and Q4 are almost exclusivelyimplemented using MOSFETs. However, other switches, such as powertransistors, IGBTs (“insulated gate bipolar transistors”), GTOs (“gateturn off” thyristors), etc, can also be used. This is again true for allthe primary switches described herein.

[0192] Q5 and Q6 are each operating in synchronous rectifier mode, whichmeans that under normal operating condition, the current will flow fromthe source terminal to drain terminal. It is noted that Q5 and Q6 can bereplaced by diodes. However, the efficiency will be reduced for VRMapplication because the loss will be much higher for diodes.

[0193] Transformer T1 can be implemented, for example, using aconventional wire wound transformer. It can also be implemented, forexample, using a planar transformer. It is noted that for VRMapplication, the planar transformer will be preferred because it canreduce the power loss and reduce the cost. This is true for all thetransformers described herein.

[0194] The inductors L1, L2 can be implemented, for example, using offthe shelf wire wound inductor. It can also be implemented, for example,using planar inductor. This is true for all the inductors describedherein.

[0195] For a typical application of a converter 301, the input voltageis 12V (which can be changed from 10.8V to 13.2V) and the output voltageis 1.5V (which can be changed from 0.8V to 1.6V). The turns ratio of thetransformer, defined as the ratio of secondary turn, Ns, and primaryturn, Np, N=Ns/Np, is selected as 0.667 (or Ns=2, Np=3). Then therequired duty cycle for Q1, defined as the ratio of on time of Q1,TonQ1, over the half switching period of Q1, 0.5*Ts, D=TonQ1/(0.5 * Ts)is about 0.5, when the loss of the converter 301 is considered. When theduty cycle is around 0.5, the performance of the converter is optimized.

[0196] The non-isolated full-bridge DC converter 1501, as shown in FIG.15, has the advantage of higher conversion efficiency. This is becausethe primary and secondary sides (high side circuit and rectifiercircuit) are not isolated;, i.e., the primary and secondary sides arecoupled directly. This will increase the efficiency, or equivalently,reduce the power loss during conversion. This will significantly improvethe life of the converter 1501, as the junction temperature ofsemiconductors is reduced. In addition, this will make the convertersize smaller. It can also simplify the mechanical design of a computermotherboard, which can reduce the cost.

[0197] As primary current goes directly to the load, this can reduce thecurrent stress of the secondary synchronous rectifiers Q5, Q6. Thecurrent ripple in the two output inductors L1, L2 will be cancelled byeach other, which reduces the output current ripple significantly.Smaller current ripple means it is possible to select a smaller outputcapacitor Co. A smaller output capacitor Co and, possibly, smallerinductor L can provide the converter 1501 with faster dynamic response.

[0198] The operation of the full-bridge converter 1501 can more easilybe understood if it is assumed that all the components are ideal. Inthis analysis, the gate drive signals Vgs1-Vgs6 for Q1, Q2, Q,3 Q4, Q5,and Q6 shown in FIG. 19 are used. There are four operating periods. Theoperation can be explained by using the equivalent circuits shown inFIG. 16, FIG. 17, and FIG. 18, and waveforms shown in FIG. 19. It shouldbe noted that other gate drive schemes can also be used to drive Q1, Q2,Q3, Q4, Q5, and Q6. One such example is the phase shifted PWM gate drivescheme, as shown in FIG. 20.

[0199] Referring to FIG. 19, the main waveforms for the full-bridgeconverter 1501 are:

[0200] 1. Vgs1 to Vgs6 are the gate signals of the 6 switches.

[0201] 2. Iin is the input current. IL1 and IL2 are the current in L1and L2.

[0202] 3. IQ5 and IQ6 are the current in switches Q5 and Q6.

[0203] 4. VL1 and VL2 are the voltage across L1 and L2.

[0204] Referring to FIG. 19, in Interval 1: From time t0 to t1, Q1, Q4and Q6 are on, resulting in the equivalent circuit 1601 shown in FIG.16. The input current Iin flows through Q1 and Q4 to the load side. Theinductor current IL1 also flows into the load side. Current in inductorL1 rises and current in L2 is falling.

[0205] Again referring to FIG. 19, at Interval 2: from time t1 to t2, Q1and Q4 are turned off and Q5 is turned on, resulting in the equivalentcircuit of FIG. 17. Therefore, Q1, Q2, Q3 and Q4 are off and Q5 and Q6are on. At this time, the input current is zero and the current in L1and L2 is falling. The energy stored in L1 and L2 is released to theload. The transformer secondary winding is shorted. The equivalentcircuit 1701 is shown in FIG. 17.

[0206] Again referring to FIG. 19, at Interval 3: from time t2 to timet3, Q2 and Q3 are turned on and Q6 is turned off. The on devices forthis interval are Q2, Q3 and Q5. The input current flows through Q2 andQ3 to the load, inductor current in L2 is rising and the current in L1is falling. The equivalent circuit 1801 is shown in FIG. 18.

[0207] Again referring to FIG. 19, at Interval 4: from time t3 to t4, Q2and Q3 are turned off and Q6 is turned on again. Q1, Q2, Q3 and Q4 areoff and Q5, Q6 are on. This stage is the same as interval 2.

[0208] At this time, the input current Iin is zero, and current IL1 andIL2 in L1 and L2 are falling. The transformer secondary winding T1B isshorted. The equivalent circuit 1701 is shown in FIG. 17.

[0209] In the above analysis, the regular PWM gate drive scheme as shownin FIG. 19 is used. A phase-shift PWM gate drive scheme is shown in FIG.20.

[0210] The main difference between the regular PWM control andphase-shift PWM control is that the latter introduces one operatinginterval when both Q1 and Q3 are on. During this period, the transformerprimary winding T1A is shorted. One advantage of the phase-shifted PWMgate drive scheme is that zero voltage switching (“ZVS”) for Q1, Q2, Q3and Q4 can be achieved by the leakage inductance of the transformer T1and by the load current. This can reduce the switching loss of Q1, Q2,Q3 and Q4. In addition, it reduces the core loss of the transformer T1by reducing the peak to peak flux density.

[0211] Based on the above analysis, some equations for the non-isolatedfull-bridge DC converter 1501 under ideal conditions will be discussed.

[0212] Relationship of output voltage Vout and input voltage Vin isshown in the following equation:

Vout=Vin*N*D/(2+N*D), N=Ns/Np  (1)

[0213] In the above equation, D is the total duty cycle and defined byD=2 * Ton/Ts, where Ton is the on time of Q1 and Ts is the switchingperiod of Q1. Ns is the turns for transformer secondary winding T1B andNp is the turns for transformer primary winding T1A. N is the turnsratio. This equation can be used to determine the turns ratio for agiven application.

[0214] The relationship between the input current Iin and output currentIo is shown in the following equation: $\begin{matrix}{I_{in\_ avg} = {I_{o\_ avg}\frac{ND}{{ND} + 2}}} & (2)\end{matrix}$

[0215] The above equation can be used to determine the input currentrequirement. It can also be used to calculate the rms (root mean square)current of MOSFETs Q1, Q2, Q3, and Q4.

[0216] The voltage stress of primary MOSFETs Q1, Q2, Q3, and Q4 is givenby the following equation.

V _(PMOSFET) =Vin−Vout  (3)

[0217] This equation is used to select MOSFETs with sufficient voltagerating. For example, for a 12V input, a MOSFET with 15V voltage ratingcan be used.

[0218] The rms current in primary MOSFETs Q1, Q2, Q3, and Q4 is given bythe following equation: $\begin{matrix}{I_{PMOSFET\_ RMS} = {\frac{I_{in}}{\sqrt{D}}\frac{\sqrt{2}}{2}}} & (4)\end{matrix}$

[0219] The above equation can be used to select MOSFETs with propercurrent rating. It can also be used to calculate the conduction loss ofthe MOSFETs.

[0220] The current ripple in the output inductors L1, L2 is given in thefollowing equation. $\begin{matrix}{{\Delta \quad I_{L}} = {\frac{V_{o}}{L}\left( {1 - {D/2}} \right){Ts}}} & (7)\end{matrix}$

[0221] In the above equation, L is the inductor value. This equation isused to calculate the peak inductor current, which is required tocalculate the switching loss, and design the inductors L1, L2.

[0222] The average current in the output inductors L1, L2 is given inthe following equation: $\begin{matrix}{I_{Lavg} = {\frac{\left( {I_{o} - I_{inavg}} \right)}{2} = {\frac{I_{o}}{2}\left( \frac{2}{{ND} + 2} \right)}}} & (8)\end{matrix}$

[0223] The above equation is used to calculate the rms current for Q5and Q6, as well as to design the inductors L1, L2.

[0224] The rms current through each synchronous rectifier Q5, Q6 isgiven by the following equation: $\begin{matrix}{I_{syn\_ RMS} = \sqrt{{\left( {1 - D} \right)I_{Lavg}^{2}} + {\frac{D}{2}\left( {I_{L1avg} + I_{L2avg}} \right)^{2}}}} & (9)\end{matrix}$

[0225] The above equation can be used to select the synchronousrectifier MOSFETs Q5 and Q6.

[0226] The above equations can be used to design the full-bridgeconverter 1501.

[0227] The above section describes in detail the operation and someadvantages of the non-isolated full-bridge DC converter 1501 under theassumption that all the components are ideal. It should be noted thatunder actual condition, the operation will be a little bit differentfrom the above analysis as will be evident to those skilled in the artusing this description.

[0228] It is also noted that by using the phase-shift PWM gate drivescheme of FIG. 20 for Q1, Q2, Q3 and Q4, zero voltage switching for theprimary MOSFETs Q1, Q2, Q3, Q4 can be achieved. This is beneficial inreducing the switching loss and/or increasing the switching frequency.

[0229] The previous section discusses in detail the basic non-isolatedfull-bridge DC converter 1501. It should be noted that when differentrectifier circuits 307 are used, other types of non-isolated full-bridgeDC converter can be derived. In this section, two alternativeembodiments of a non-isolated full-bridge DC converter will bedescribed. Their operation is similar to the original embodiment 1501 asshown in FIG. 15. As will be evident to those skilled in the art, otherembodiments can also be derived using similar methods.

[0230] Referring to FIG. 21, another embodiment 2101 of a non-isolatedfull-bridge converter is shown. In this embodiment, the rectifiercircuit includes two secondary windings of the transformer. Diode D1 andD2 are used to convert the bi-directional voltage across the secondarywindings of the transformer T1 into one-direction pulsating voltage.Only one inductor L is used.

[0231] Referring to FIG. 22, when the diodes D1, D2 in the circuit 1501shown in FIG. 15 are replaced by synchronous rectifiers Q5, Q6 in orderto reduce the power loss in low output voltage applications, the circuit2201 is obtained. The position of Q5, Q6 is interchanged with thesecondary winding T1B, T1C in order to simplify the requirement for thegate drive of Q5 and Q6.

[0232] In order to increase the current carrying capability or to reducepower loss in a rectifier circuit, two or more rectifier sections can beconnected in parallel.

[0233] Referring to FIG. 23, one such circuit 2301, which is based oncircuit 1501 in FIG. 15, is derived by using two rectifier sections2303, 2305 (each section forming its own rectifier circuit). The tworectifier sections 2303, 2305 are connected in parallel. One rectifiersection 2303 consists of Q5, Q6, L1, L2 and one secondary winding T1C.The other rectifier section 2305 consists of Q7, Q8, L3, L4 and onetransformer secondary winding T1B. Transformer T1 has one primarywinding T1A (same as in FIG. 15), and has two secondary windings T1B,T1C. These three windings T1A, T1B, T1C are coupled together throughmagnetic core of transformer T1.

[0234] Using this method the output load current is shared in the twosecondary windings T1B, T1C. As we know the conduction losses are I²R,if the resistance is the same, and the current becomes half, theconduction losses will reduce four times. This is more effective thanusing two MOSFETs in parallel to reduce conduction losses when parasiticparameters are considered. In other words, two secondary windings T1B,T1C arrangement will ensure better current sharing in rectifier circuit2301.

[0235] The major advantage of this arrangement is the reduction of theconduction loss for inductors L1-L4 and synchronous rectifiers Q5, Q6,Q7, Q8. When the transformer T1 is implemented by planar transformer,the cost of adding another winding is zero. The gate drive of Q7 is thesame as that of Q5 and the gate drive of Q8 is same as that of Q6. Whenthe same MOSFETs are used, the conduction loss for the rectifier circuitcan be cut by half. That can increase the efficiency by about 3%.

[0236] In cost-sensitive applications, lower cost MOSFETs (whichnormally have higher on resistance) can be used, and the totalconduction loss will be similar to that of FIG. 15, while the cost offour lower cost MOSFETs may well be lower than the cost of two premiumMOSFETs.

[0237] Referring to FIG. 24, another rectifier circuit embodiment 2401including two rectifier sections 2403, 2405 is shown. In this figure,the two rectifier sections 2403, 2405 are each the same as the rectifiercircuit used in FIG. 21, and they are connected in parallel. Rectifiersection 2403 consists of secondary windings T1B, T1C, diodes D1 and D2,and inductor L1. Rectifier section 2405 consists of secondary windingsT1D, T1E, diodes D3 and D4, and inductor L2. It is noted that all thefive transformer windings, T1A, T1B, T1C, T1D and T1E, are coupled tothe same magnetic core of transformer T1.

[0238] Referring to FIG. 25A, another rectifier circuit embodiment 2501including two rectifier sections 2503, 2505 is shown. In this figure,the two rectifier sections 2503, 2505 are each the same as the rectifiercircuit used in FIG. 22, and they are connected in parallel. Rectifiersection 2503 consists of secondary windings T1B, T1C, MOSFETs Q5 and Q6,and inductor L1. Rectifier section 2505 consists of secondary windingsT1D, T1E, MOSFETs Q7 and Q8, and inductor L2. It is noted that all thefive transformer windings, T1A, T1B, T1C, T1D and T1E, are coupled tosame magnetic core of transformer T1.

[0239] It is noted that in the above discussion of FIGS. 23-25A, thenumber of rectifier circuits is limited to two. In actualimplementation, three or more rectifier sections can also be used toshare the load current in order to reduce the conduction loss in therectifier circuit.

[0240] It should be noted that in actual implementation, adding moresecondary windings does not necessarily increase the transformer costwhen a planar magnetic structure is used; however, every time asecondary winding is added, more inductors are also needed.

[0241] In order to increase the current carrying capability, two or moreidentical DC converters can be connected in parallel with respectiveinputs and outputs connected together. Interleaving is a technology thatcontrols the turn on instant of the switches in different DC convertersso that the input current ripple and output current ripple can besignificantly reduced. For example, the turn on instant of one MOSFET inthe second converter is delayed with respect to the same MOSFET in thefirst converter so that the input currents of these two converters are180 degrees out of phase. This will result in significant reduction ofthe input current ripple and, therefore, reduction of the size of theinput filter. By interleaving, the output current ripple of eachconverter is also 180 degrees out of phase. This will also result insignificant reduction of the output current ripple and, therefore, thesize of the output filter. It should be noted that a smaller inputfilter and output filter is very beneficial in improving the transientresponse of the converter.

[0242] One extension of the circuit 301 shown in FIG. 3 is the two-phaseinterleaved circuit 25A01, as shown in FIG. 25B. Referring to FIG. 25B,two identical high side circuits are connected in parallel and twoidentical rectifier circuits are connected in parallel also. The gatedrive signals to the high side circuit 1 and high side circuit 2 arephase shifted. The gate drive signals to the rectifier circuit 1 andrectifier circuit 2 are also phase shifted. This arrangement can achieveall the benefits of interleaving.

[0243] Interleaving can improve the performance of DC converterssignificantly. The improvement of the interleaf is even more significantfor two-phase interleaving when the duty cycle in each converter isaround 50%. It is noted that the steady state duty cycle of theconverter is around 50%. Therefore, the benefit of interleaving is verysignificant as compared with a conventional interleaved Buck converter.

[0244] It should be noted that FIG. 25B only shows the implementation oftwo-phase interleaved converter. Three-phase or even more-phaseinterleaving can also be derived in same way as will be evident to thoseskilled in the art using the principles described herein.

[0245] Several ways to implement interleaving for non-isolatedfull-bridge DC converters are described in the following paragraphs.Their operation is discussed briefly.

[0246] Referring to FIG. 26, a two-phase interleaved non-isolatedfull-bridge DC converter 2601 is shown. In the figure, top non-isolatedfull-bridge DC converter 2603 includes Q1, Q2, Q3, Q4, Q5, Q6, T1, L1and L2. Bottom non-isolated full-bridge DC converter 2605 includes Q7,Q8, Q9, Q10, Q11, Q12, T2, L3 and L4. Capacitor Co belongs to bothconverters 2603, 2605. Co could be implemented as separate capacitorsfor each converter 2603, 2605. Due to the interleaving the value of Cocan be reduced. In addition, these two power stages operate in 90 degreephase shift.

[0247] Referring to FIG. 27, the main waveforms of the two-phaseinterleaved non-isolated full-bridge DC converter 2601 are shown. Thegate drive signals for Q1, Q2, Q3, Q4, Q5, and Q6 are the same as thosein FIG. 19; however, the gate drive signal for Q10 is delayed by aquarter of Ts (0.25 * Ts or 90 degrees) with respect to Q1. Similarly,the gate drive signal for Q9 is delayed by a quarter of Ts or 90 degreeswith respect to Q2. The gate drive signal for Q12 is delayed by 90degrees with respect to Q3. The gate drive signal for Q11 is delayed by90 degrees with respect to Q4. The gate drive signal for Q7 is delayedby 90 degrees with respect to Q5. The gate drive signal for Q8 isdelayed by 90 degrees with respect to Q6. It can be observed from FIG.27 that the ripple current through the input capacitor and outputcapacitor is significantly reduced. In addition, the ripple currentfrequency is four times the switching frequency. This fact is verybeneficial for reducing the capacitor size and therefore, improving thedynamic response.

[0248] In FIG. 27, a duty cycle of 40% is assumed. Referring to FIG. 28,the ripple cancellation effect is even more significant when the dutycycle is 50%. In this case, the input current is continuous with onlyinductor ripple present. The input current is always higher than zero.

[0249] Referring to FIG. 29, the waveforms of a two-phase interleavedfull-bridge converter 2601 when duty cycle is 60% are shown. Again,significant ripple current reduction is achieved.

[0250] From the above analysis, it is demonstrated that by using theinterleaved method, the input current ripple and output current rippleof the two-phase interleaved non-isolated full-bridge DC converter 2601can be reduced significantly. In addition, the current ripple frequencyis four times the switching frequency. This makes it possible to selecta smaller output inductor and a smaller output capacitor. It is notedthat a small output capacitor and a small output inductor can helpachieve better dynamic response of the power converter 2601. Inaddition, the cost of the converter 2601 can be reduced.

[0251] Referring to FIG. 30, a two-phase interleaved non-isolatedfull-bridge DC converter 3001 with shared switch is shown. As comparedwith the interleaved non-isolated full-bridge DC converter 2601, onlysix MOSFETs Q1-Q6 are used in the high side circuit 3003, as the numberof primary switches is reduced, while the topology in FIG. 30 canrealize the same function as the topology in FIG. 26.

[0252] Referring to FIG. 31, a special gate drive scheme should be used.One embodiment of the gate drive signal arrangement for all the switchesin the above circuit is shown. Q2 and Q5 always conduct for half theswitching period out of phase, which means Q2 conducts for the firsthalf cycle and Q5 conducts for the second half cycle, as shown in thefigure. Q1 and Q3 phase shift from each other. It is noted that in thecontrol logic shown in FIG. 31, the move direction of Q1 and Q3 aredifferent from conventional phase shift control. Q1 is turned on at thesame time as Q5 is turned on. Q1 is turned off after on time TQ1on. Q3should be turned off when Q5 is turned off. The turn on time of Q3 iscontrolled to achieve the required conduction time, TQ3on. Thisrequirement is illustrated by the arrow in FIG. 31. Q4 and Q6 work inthe same way as Q3 and Q5. For synchronous rectifiers, Q7 is driven bythe complementary signal of Q6. Q8 is driven by the complementary signalof Q3. Q9 is driven by the complementary signal of Q1. Q10 is driven bythe complementary signal of Q4, as shown in FIG. 31.

[0253] Referring to FIG. 32, the input current and output current areshown, as well as the gate drive signals for the two-phase interleavednon-isolated full-bridge DC converter 3001 with high side circuit sharedswitch when the duty cycle is 40%. In the waveform, the current ripplefor the inductor is neglected. It can be observed that the AC componentof input current and output current is significantly reduced.

[0254] Referring to FIG. 33, the impact of the ripple cancellation iseven more significant when the duty cycle is 50%. When the duty cycle is50%, the AC component of the input current and output current iscomplete cancelled. The input and output current is a pure DC value. Inreality, only inductor ripple current appears at the input and output.

[0255] Referring to FIG. 34, the input current and output currentwaveforms when the duty cycle is 60% are shown. Again, significantripple reduction is achieved.

[0256] Referring to FIG. 35A, another two-phase interleaved non-isolatedfull-bridge DC converter 3501 with high side circuit shared switches andrectifier circuit shared switches is shown. In this converter, thenumber of rectifier circuit switches is reduced from four to three. Thenumber of inductors is reduced to three also. This simplifies thecontrol logic. At the same time, it can achieve all the functions thetopology in FIG. 30 can achieve. It is noted that the rectifier circuithas fewer components; however, this topology will have slightly moreconduction loss when compared with converter 3001 in FIG. 30.

[0257] Similar to two-phase interleaving to reduce the input and outputcurrent ripple, three-phase interleaving can be achieved by using threenon-isolated full-bridge DC converter connected in parallel. FIG. 35Bshows the schematic diagram of a three-phase interleaved non-isolatedfull-bridge converter. In this circuit, 12 high side switches are used.

[0258] The technique to share the high side switch can also be used inthe case of three-phase interleaved non-isolated full-bridge DCconverter. Referring to FIG. 36, again, similar to two-phaseinterleaving with shared high side switch, a three-phase interleavednon-isolated full-bridge DC converter with shared high side switch isshown. Another pair of switches and one transformer primary winding isadded. The circuit shown in FIG. 36 requires only eight high sideswitches and it can achieve the same performance as a three-phaseconverter without shared switches, as shown in FIG. 35B. In same way,four- or more-phase interleaved non-isolated DC converters with highside circuit shared switches can be derived. They are not illustratedhere.

[0259] Referring to FIG. 37A and FIG. 37B, a special gate drive schemeis needed. One embodiment of the gate drive scheme is shown for thethree-phase full-bridge converter 3601 with 8 primary switches. Thistopology can realize the same function as the basic three-phaseinterleaved non-isolated full-bridge converter shown in FIG. 35B. Inthis circuit, Q1 and Q2 conduct out of phase. Q3 and Q4 conduct out ofphase also. Q3 and Q4 phase shift according to Q1 and Q2. Q5 turns onwhen Q4 turns on, Q6 turns on when Q3 turns on, and Q6 and Q5 onlychange their turn off time. Q7 turns off when Q2 turns off and Q8 turnsoff when Q1 turns off. Q7, Q8 only change their turn on time. Thistopology will have the smallest input and output current ripple when theduty cycle is 33.3%. The waveform is shown in FIG. 37B.

[0260] When compared with the two-phase circuit 3001 shown in FIG. 30,this topology adds two more high side circuit switches, anothertransformer, and two rectifier circuit switches. One can add moreswitches in the high side and rectifier circuits to form a four-phaseshift or even more-phase shift converter, and the rectifier circuitswitches can also be shared as shown in FIG. 35A. In this way, thefrequency of input and output current can be increased, and the currentripple of the input and output current reduced. Current stress of theswitches can also be reduced. This has advantages in high power and highcurrent applications.

[0261] Referring to FIG. 38A, another three-phase interleavednon-isolated full-bridge converter 3801 is shown. A third primarywinding T3A is added to the high side circuit of converter 3001 inaddition to other primary windings T1A and T2A. Therefore, only six highside switches are needed.

[0262] One gate drive scheme for the converter 3801 is provided in FIG.38B. Referring to FIG. 39, a circuit topology for three-phaseinterleaved non-isolated full-bridge DC converter 3901 with rectifiercircuit shared switches is shown.

[0263] In these topologies, six high side switches form three powerstages and phase shift from each other. The output power is sharedbetween the three power stages. The current ripple of the three powerstages when added together is three times switching frequency and thecurrent ripple of the three power stages can cancel each other. Thesetopologies have advantages in high current applications.

[0264] The topology in FIG. 39 has fewer switches in the rectifiercircuit than the topology in FIG. 38A, while it can realize the samefunction as the topology in FIG. 38A. This makes the control logicsimpler. This topology will have more conduction losses when compared tothe topology in FIG. 38A.

[0265] The above section describes several non-isolated DC converterswhen the circuit 401 shown in FIG. 4 is used as the high side circuit305. Detailed analysis has been given for a basic non-isolatedfull-bridge DC converter 1501 as shown in FIG. 15. As will be evident tothose skilled in the art using the principles described herein, theanalysis of other converters described herein can be done in a similarway.

[0266] It should be noted that the above analysis has not explored allthe combinations of the new circuits. A person skilled in the art canderive other types of circuit configuration using the same methodologypresented in this disclosure. The following section describes theoperation of the non-isolated DC converters when the high side circuit(as shown in FIG. 3) is derived using two MOSFETs as switches, onetransformer primary winding and two capacitors as a voltage divider.Several improved embodiments based on this arrangement are describedbelow.

[0267] Referring to FIG. 40, a non-isolated half-bridge DC converter4001 is shown. It has similar characteristics to basic non-isolatedfull-bridge DC converter 1501 as shown in FIG. 15. As the input currentgoes directly to the load, the current stress of the synchronousrectifiers is reduced.

[0268] By selecting the turns ratio N properly, the duty cycle D of theconverter can be around 50%, which is beneficial for the performance ofthe converter 4001.

[0269] The operation of the basic non-isolated half-bridge DC converter4001 shown in FIG. 40 will now be described.

[0270] Referring to FIG. 44, the main waveforms for the non-isolatedhalf-bridge DC converter 4001 are shown, where:

[0271] 1. Vgs1 to Vgs4 are the gate signals of the four switches: twohigh side switches Q1, Q2 and two rectifier switches Q3, Q4.

[0272] 2. Iin is the input current, and IL1 and IL2 are the current inL1 and L2.

[0273] 3. IQ3 and IQ4 are the current in switches Q3 and Q4, VL1 and VL2are the voltage across L1 and L2.

[0274] In Interval 1: from t0 to t1, Q1, Q3 are on. The input currentflows through Q1 and the transformer primary winding to the load. Theinductor current also flows into the load side. The current in inductorL1 rises and the current in L2 is falling. The equivalent circuit 4101is shown in FIG. 41.

[0275] At Interval 2: from t1 to t2, Q1 is turned off and Q4 is turnedon. Therefore, Q1, Q2 are off and Q3 and Q4 are on. During thisinterval, the input current is zero. The inductor current and outputcapacitor provide the load current. The current in L1 and L2 arefalling. The energy stored in L1 and L2 is released to the load. Thetransformer secondary winding is shorted. The equivalent circuit 4201 isshown in FIG. 42.

[0276] At Interval 3: from t2 to t3, Q2 is turned on and Q3 is turnedoff. The on devices for this interval are Q2 and Q4. The input currentflows through Q2 and the transformer primary winding to the load. Theinductor current also provides current to the load. The inductor currentin L2 is rising and the current in L1 is falling. The equivalent circuit4301 is shown in FIG. 43.

[0277] At Interval 4: from t3 to t4, Q2 is turned off and Q3 is turnedon again. Q1, Q2, are off and Q3, Q4 are on. This stage is same asinterval 2. During this interval, the inductor current and outputcapacitor provide the load current. The input current is zero. Thecurrent in L1 and L2 is falling. The transformer secondary winding isshorted. The equivalent circuit 4201 is in FIG. 42

[0278] From the above analysis, the input and output voltage relationcan be derived as:

Vout=N*D*Vin/(4+2*D*N)

[0279] Where D is the duty cycle, defined as TonQ1/(0.5 * Ts), N=Ns/Npand Vin is the input voltage.

[0280] The previous section discusses in detail a basic non-isolatedhalf-bridge DC converter 4001. When different rectifier circuits areused, other types of non-isolated half-bridge DC converter can bederived. In this section, two alternative embodiments of thenon-isolated half-bridge DC converter are shown. Their operation issimilar to the original embodiment as shown in FIG. 40. Otherembodiments can also be derived using same method.

[0281] Referring to FIG. 45, in converter 4501 a high side circuitconsists of two MOSFETs Q1, Q2 and two capacitors C1, C2. Otherwise, theconverter 4501 is the same as the converter 2101 of FIG. 21, as therectifier circuit is same.

[0282] Referring to FIG. 46, synchronous rectifiers Q3, Q4 in converter4601 replace the diodes D1, D2 in the converter 4501 shown in FIG. 45 inorder to reduce the power loss in low output voltage applications. Theposition of Q3, Q4 is interchanged with the secondary windings in orderto simplify the requirement for the gate drive of Q3 and Q4.

[0283] In order to increase the current carrying capability or to reducepower loss in rectifier circuits, two or more rectifier sections (eachrectifier section including its own rectifying circuit) can be connectedin parallel.

[0284] Referring to FIG. 47, a non-isolated half-bridge DC converter4601 with a rectifier circuit 4703 having two rectifier sections 4705,4707 is shown. Converter 4701 has two secondary windings T1B, T1Ccoupled with the primary winding T1A. The load current is shared in thetwo secondary windings T1B, T1C. As the conduction losses is I²R, if theresistance is the same, and the current is halved, the overallconduction losses are reduced by two times, this is more effective thanparallel switches to reduce conduction losses when the impact of theparasitic components are considered. Referring again to FIG. 47, thereare two secondary windings T1B, T1C in parallel. Three or four or evenmore secondary windings can be paralleled to share the load current.Every time one secondary winding is added, two inductors also need beadded.

[0285] Two or more other types of rectifier circuits, such as thoseshown in FIG. 13 and FIG. 14, can also be connected in parallel toincrease the output current carrying capability and/or reduce theconduction loss at the rectifier circuits. Detailed circuit diagrams arenot shown in this description, but would be evident to one skilled inthe art employing the principles described herein.

[0286] Multi-phase interleaving technology can also be used innon-isolated half-bridge converters to reduce input and output currentripple and to improve dynamic response.

[0287] Referring to FIG. 48, a two-phase interleaved half-bridgeconverter 4801 is shown. The output power is shared between the twopower phases of the topologies. One phase consists of Q1 and Q2 as highside switches, Q5 and Q6 as synchronous rectifier switches, T1A as theprimary winding, T1B as the secondary winding, and L1 and L2 as theoutput inductors. Another phase consists of Q3 and Q4 as high sideswitches, Q7, and Q8 as synchronous rectifier switches, T2A as theprimary winding, T2B as the secondary winding, and L3 and L4 as theoutput inductors. The capacitors C1 and C2 are shared by the two phases.

[0288] Referring to FIG. 49, gate drive signals Vgs1-Vgs8 and input andoutput current Iin, Iout waveforms are shown. Q1 and Q3 are phaseshifted by 90 degrees from each other. This means that Q3 is alwaysturned on a quarter switching period (0.25 * Ts) after Q1 is turned on.Similarly, Q4 is always turned on a quarter switching period after Q2 isturned on. The gate drive for Q5 is similar to the basic non-isolatedhalf-bridge DC converter as shown in FIG. 40. Output current ripple ofthese two phases can be reduced significantly. The frequency of theripple current is two times the switching frequency. The input andoutput ripple current of the two phases also can cancel each other. Thismakes the selection of a smaller output inductor and a smaller outputcapacitor possible. It is noted that a small output capacitor and asmaller output inductor can help improve the dynamic response of thepower converter.

[0289] Q1, Q2 and Q3, Q4 form two power phases that are phase shifted 90degrees from each other. A duty cycle of 40% is shown. It is noted thatwhen the duty cycle is 50%, the input and output current ripple is thesmallest.

[0290]FIG. 50 shows a two-phase interleaved non-isolated half-bridge DCconverter 5001 with shared rectifier switch. This converter 5001 hasfewer switches and inductors than the converter 4801 in FIG. 48. It canachieve the same function as the converter 4801 in FIG. 48. This makesthe control logic simpler and lower cost. This converter 5001 has aslightly higher conduction loss than the converter 4801 in FIG. 48.

[0291] The above section describes several non-isolated DC converterswhen the circuit 601 shown in FIG. 6 is used as the high side circuit305. Detailed analysis has been given for basic non-isolated half-bridgeDC converter 4001 as shown in FIG. 40. The analysis of other convertersusing the principles described herein can be done in a similar way.

[0292] The above analysis has not explored all the possible combinationsof circuits using the principles described herein to produce novelconverters. A person skilled in the art can derive other circuitconfigurations using the principles presented in this description.

[0293] Referring to FIG. 51, when the circuit 801 shown in FIG. 8 isused as the high side circuit 305 and when the circuit 1101 shown inFIG. 11 is used as the rectifier circuit 307, a non-isolated forward DCconverter 5101 is derived. In the converter 5101, transformer T1 has oneprimary winding T1A and one secondary winding T1B.

[0294] By selecting the turns ratio N properly, the duty cycle D of theconverter 5101 can be around 50%, which is beneficial for theperformance of the converter 5101.

[0295] When Q1 and Q2 are on, the input current Iin goes directly to theoutput Vout through transformer primary winding T1A and Q1, Q2. Inaddition, some of the energy is transferred to the transformer secondarywinding T1B and is rectified into DC voltage by diodes D3 and D4, andfiltered by inductor L.

[0296] Referring to FIG. 52, when the circuit 801 shown in FIG. 8 isused as high side circuit 305 and the circuit 1101 shown in FIG. 11 isused as the rectifier circuit 307, another non-isolated forward DCconverter 5201 is derived. In converter 5201, synchronous rectifiers areused to reduce the power loss for low output voltage applications (suchas VRM applications).

[0297] Referring to FIG. 53, when the circuit 801 shown in FIG. 8 isused as high side circuit 305 and the circuit 1001 shown in FIG. 10 isused as the rectifier circuit 307, another non-isolated forward DCconverter 5301 is derived. In converter 5301, two synchronous rectifiersand two inductors are used.

[0298] Referring to FIG. 54, when the circuit 901 shown in FIG. 9 isused as the high side circuit 305 and when the circuit 1201 shown inFIG. 12 is used as the rectifier circuit 307, another non-isolatedforward DC converter 5401 is derived. In the circuit 5401, transformerT1 has one primary winding T1B and one secondary winding T1B. When Q1 ison, the input current goes to output directly through transformerprimary winding T1A and Q1. In addition, some of the energy istransferred to the transformer secondary winding T1B and is rectifiedinto DC voltage by diodes D1, D2 and filtered by inductor L.

[0299] Referring to FIG. 55, when the circuit 901 shown in FIG. 9 isused as high side circuit 305 and the circuit 1101 shown in FIG. 11 isused as the rectifier circuit 307, another non-isolated forward DCconverter 5501 is derived.

[0300] This converter 5501 has the advantage of lower conduction lossbecause only one power MOSFET Q1 is used in the high side. Q1 and Q3 areturned on at same time. Q2 and Q4 are turned on at same time. When Q1 ison, the input power is transferred to output via two paths. One path isfrom the transformer primary winding T1A and Q1 to the output load. Theother path is from the secondary winding T1B, Q3 and inductor L to theoutput load. When Q1 is off, Q2 is on. The transformer core is reset bythe voltage across C1. The energy stored in inductor L is alsotransferred to the output load during this period.

[0301] Referring to FIG. 56, when the circuit 901 shown in FIG. 9 isused as high side circuit 305 and the circuit 1001 shown in FIG. 10 isused as the rectifier circuit 307, another non-isolated forward DCconverter 5601 is derived. In this converter 5601, two inductors areused.

[0302]FIG. 57 shows another non-isolated forward DC converter, converter5701. In converter 5701, high side circuit 5703 consists of onetransformer primary winding T1A, one switch Q1, one reset winding T1Cand one diode D1. The rectifier circuit 5703 consists of one transformersecondary winding T1B, two synchronous rectifiers Q2 and Q3 and inductorL. Reset winding T1C and diode D1 are used to reset the core of thetransformer. Q1 and Q2 are turned on at same time. When Q1 is on, theenergy is transferred from input to output through two paths. One isfrom T1A and Q1 to the output. The other is from T1B, Q2, and inductor Lto the output. When Q1 is off, Q3 is on. The energy stored in L and Cois released to the output load. The diode D1 is turned on and the inputvoltage is applied to reset winding T1C. In this way, the transformercore is reset by Vin.

[0303] Similarly, when multiple rectifier circuits are connected inparallel, more load current can be provided. Or equivalently, conductionloss can be reduced. The circuit details can be derived using theprinciples described herein, and are not further described herein.

[0304] Similarly, multi-phase interleaved technology can also be used inall the above non-isolated forward converters shown from FIG. 51 to FIG.57 to reduce the input and output current ripple. The improvementintroduced by interleaving can be very significant because these DCconverters are operated at a duty cycle of around 50%.

[0305] Referring to FIG. 58, when two converters 5101, as shown in FIG.51, are connected in parallel, one two-phase interleaved forward DCconverter 5801 is derived. The gate drive signals to converter 5803 andconverter 5805 are interleaved to reduce the input and output currentripple.

[0306] Referring to FIG. 59, when two converters 5201, as shown in FIG.52, are connected in parallel, another two-phase interleaved forward DCconverter 5901 is derived. The gate drive signals to converter 5903 andconverter 5905 are interleaved to reduce the input and output currentripple.

[0307] Referring to FIG. 60, when two converters 5301, as shown in FIG.53, are connected in parallel, another two-phase interleaved forward DCconverter 6001 is derived. The gate drive signals to converter 6003 andconverter 6005 are interleaved to reduce the input and output currentripple.

[0308] Referring to FIG. 61, when two converters 5401, as shown in FIG.54, are connected in parallel, another two-phase interleaved forward DCconverter 6101 is derived. The gate drive signals to converter 6103 andconverter 6105 interleaved to reduce the input and output currentripple.

[0309] Referring to FIG. 62, when two converters 5501, as shown in FIG.55, are connected in parallel, another two-phase interleaved forward DCconverter 6201 is derived. The gate drive signals to converter 6203 andconverter 6205 are interleaved to reduce the input and output currentripple.

[0310] Referring to FIG. 63, when two converters 5601, as shown in FIG.56, are connected in parallel, another two-phase interleaved forward DCconverter 6301 is derived. The gate drive signals to converter 6303 andconverter 6305 are interleaved to reduce the input and output currentripple.

[0311] Other non-isolated circuits can be derived using the principlesdescribed herein and are not further described herein.

[0312] The principles for creating a new family of non-isolated DCconverters wherein the input current of the high side circuit isconnected directly to the load have been described. Several embodimentsof high side circuits have been illustrated. Several embodiments ofrectifier circuits are also illustrated.

[0313] Using different combinations of high side circuits and rectifiercircuits, new non-isolated DC converters can be derived. Some suchembodiments are shown in this description. It should be noted that othernon-isolated DC converters can be derived using the principles describedherein.

[0314] In addition, multiple rectifier circuits can be used to reducethe conduction loss for the rectifier circuits in high load currentapplications.

[0315] Interleaving technology can also be used to significantly improvethe performance of the non-isolated DC converter proposed in thisdescription. The main reason for this improvement is that all the DCconverters derived using the proposed method will operate at duty cyclesof around 50%, which is beneficial in reducing significantly the inputcurrent and output current ripple, as well as in improving theefficiency and dynamic response. Waveforms have been used to illustratethe benefit of interleaving.

[0316] It is noted that for two-phase interleaved converters describedin this specification, current sharing between the two converters shouldbe implemented. By current sharing, each converter will provide half theload current. One implementation of current sharing between circuits25A01, as shown in FIG. 25B, is illustrated in FIG. 64.

[0317] In FIG. 64, current sensing circuits 6413 and 6415 are connectedin series with the high side circuits to sense the current through eachhigh side circuit. It is noted that the current through the high sidecircuit is same as the input current to each high side circuit. Incircuit 6401, resistors Rs1, Rs2 are used as the current sensing circuit6413 and 6415. In actual implementation, other types of current sensingcircuits, such as a current sensing transformer, can also be used, aswill be evident to those skilled in the art using the principlesdescribed herein.

[0318] The advantage of placing the current sensing circuit in serieswith high side circuit is to reduce the power loss. The current throughthe high side circuit is smaller than the load current, and therefore,the power loss in the current sensing circuit is smaller.

[0319] Vs1 is the voltage across the current sensing resistor Rs1 andVs2 is the voltage across current sensing resistor Rs2. Vs1 and Vs2 arefed into the current sharing circuit 6417. The output of the currentsharing circuit 6417 is a current sharing signal, Ishare. The currentsharing signal Ishare is fed into PWM controller for converter 1 and PWMcontroller for converter 2. It should be noted that converter 1 consistsof high side circuit 1 and rectifier circuit 1. Converter 2 consists ofhigh side circuit 2 and rectifier circuit 2.

[0320] It is noted that same method can be used for multiple phasecurrent sensing scheme as will be evident to those skilled in the artusing the principles described herein.

[0321]FIG. 65 shows one implementation of a current sensing circuit fora two-phase interleaved full-bridge DC converter. Referring to FIG. 65,current sensing resistor Rs1 is connected between the source of Q2, Q4and the positive point of the output voltage. Current sensing resistorRs2 is connected between the source of Q9, Q11 and the positive point ofoutput voltage. Vs1 is the voltage across Rs1 and Vs2 is the voltageacross Rs2. Vs1 and Vs2 are fed into a current sharing circuit, which isnot shown in the figure.

[0322]FIG. 66 shows one implementation of a current sensing circuit fora two-phase interleaved half-bridge DC converter. Referring to FIG. 66,current sensing resistor Rs1 is connected between the source of Q2 andthe positive point of the output voltage. Current sensing resistor Rs2is connected between the source of Q4 and the positive point of outputvoltage. Vs1 is the voltage across Rs1 and Vs2 is the voltage acrossRs2. Vs1 and Vs2 are fed into a current sharing circuit, which is notshown in the figure.

[0323]FIG. 67 shows one implementation of a current sensing circuit fora two-phase interleaved forward DC converter. Referring to FIG. 67,current sensing resistor Rs1 is connected between the source of Q1 andthe positive point of the output voltage. Current sensing resistor Rs2is connected between the source of Q6 and the positive point of outputvoltage. Vs1 is the voltage across Rs1 and Vs2 is the voltage acrossRs2. Vs1 and Vs2 are fed into a current sharing circuit, which is notshown in the figure.

[0324] The above circuits, FIG. 65 to FIG. 67, illustrate examples ofhow to implement a current sensing circuit in the converter circuitsderived in this specification. The figures show the implementation fortwo-phase interleaved full-bridge, half-bridge and forward DCconverters, using a resistor as a current sensing component. It will beevident to those skilled in the art using the principles describedherein how to implement current sensing circuits in other topologiesproposed in this specification.

[0325] Referring to FIG. 68A, a new family of non-isolated resonant DCconverter 9601 is shown. Similar to the converter 301, converter 9601has a high side circuit 9602, a rectifier circuit 9604 and an outputcapacitor Co. The high side circuit 9602 is similar to high side circuit305; however, in addition the converter 9601 has a resonant tankoperating in the high side circuit 9602. The resonant tank filters outvery high frequency components of energy waveforms in the high sidecircuit 9602, thus smoothing the waveforms in the high side circuit 9602to transform the waveforms from an almost square wave to aquasi-sinusoidal waveform. This reduces switching losses in theauxiliary section of the thigh side circuit 9602. The timing andoperation of the converter 9601 is similar to that of the converterspreviously described herein.

[0326] A number of non-isolated full-bridge parallel resonant converterembodiments of the converter 9601 will now be described. Referring toFIG. 68B, a non-isolated full-bridge parallel resonant converter 6801has 8 MOSFETs (Q1 to Q8), one transformer (T1), with one primary windingand one secondary winding. A resonant tank consists of one inductor (L1)and one capacitor (C1), the capacitor C1 is paralleled with the primarywinding of T1 then placed in series with inductor L1. One outputinductor (L2) and one output capacitor (Cout) are used.

[0327] As with previously described converters, the converter 6801 hastwo circuits: a high side circuit having a transformer T1 primarywinding, an auxiliary section (Q1 to Q4) and a resonant tank (L1, C1),and a rectifier circuit 6802 with Q5 to Q8, a transformer T1 secondarywinding, and L2 and Cout, forming a full-bridge rectifier circuit 6802.

[0328] Q1 to Q4 are primary switches. They can be implemented by, forexample, IRF7467 from International Rectifier. Q5 to Q8 are synchronousrectifier switches, they can be implemented by, for example, IRLR8103from International Rectifier or be replaced by diodes. Other switchessuch as IGBT or GTO can also be used for Q1 to Q8.

[0329] Those skilled in the art will appreciate that switching frequencycontrol can be used for all switches in the converters 9601 to regulatethe output voltage across the load. Similarly, phase shift control canbe used for high side switches in any full-bridge high side circuit (forexample, the high side circuits in FIGS. 68B through 81 as will bedescribed) in the converter 9601. Switching frequency control and phaseshift control may be used alone or in combination. The implementation ofswitching frequency control and phase shift control in high sidecircuits is known in the art, see for example [1] R. L. Steigerwald,“High frequency resonant transistor DC-DC converters”, IEEE Transactionson Industrial Electronics, Vol. 31, pp. 181-181, May 1984; [2] F. S.Tsai, P. Matera, and F. C. Lee, “Constant-frequency, clamped-moderesonant converters”, IEEE Transactions on Power Electronics, Vol. 3,No. 4, pp. 460-473, October 1988; and [3] Y. F. Liu and P. C. Sen,“Source reactance lossless switch (SRLS) for soft-switching converterswith constant switching frequency”, IEEE Transactions on Circuit andSystems—I, Fundamental Theory and Applications, Vol. 43, No.4, pp.301-312, April 1996.

[0330] Referring to FIG. 69, an alternate full-bridge parallel resonantconverter 6901 is shown. Its high side circuit is the same as thetopology shown in FIG. 68B, but the rectifier circuit is changed from afull-bridge rectifier in FIG. 68B to a current doubler 6902. The currentdoubler 6902 has two MOSFETs (Q5, Q6), two output inductors (L2, L3),one secondary winding of the transformer T1 and an output capacitor(Cout).

[0331] Q5 and Q6 are synchronous rectifier switches. They can beimplemented by, for example, IRLR8103 from International Rectifier or bereplaced by diodes. Other switches such as IGBT or GTO can also be usedfor Q1 to Q6.

[0332] Referring to FIG. 70, a further alternate full-bridge parallelresonant converter 7001 is shown. Its high side is the same as thetopology shown in FIG. 68B, but the rectifier circuit is changed from afull-bridge rectifier in FIG. 68B to a center tapped transformerrectifier 7002. The center tapped transformer rectifier 7002 has twoMOSFETs (Q5, Q6), one output inductor (L2), and two secondary windingsof the transformer T1, and an output capacitor (Cout).

[0333] Q5 and Q6 are synchronous rectifier switches. They can beimplemented by, for example, IRLR8103 from International Rectifier or bereplaced by diodes. Other switches such as IGBT or GTO can also be usedfor Q1 to Q6.

[0334] Referring to FIG. 71, a further alternate full-bridge parallelresonant converter 7101 is shown. Its high side circuit is the same asthe topology shown in FIG. 68B, but the rectifier circuit is changedfrom a full-bridge rectifier in FIG. 68B to a phase control rectifier7102. The phase control rectifier 7102 consist of four MOSFETs (Q5 toQ8), three diodes (D1 to D3), three inductors (L2 L3 and L4), onesecondary winding of the transformer T1 and an output capacitor (Cout).

[0335] D1 to D3 can also be replaced by MOSFETs to reduce conductionloss. Q5 and Q8 are synchronous rectifier switches. They can beimplemented by, for example, IRLR8103 from International Rectifier.Other switches such as IGBT or GTO can also be used for Q1 to Q8.

[0336] Those skilled in the art will appreciate that all full-bridgerectifier circuits in converters 9601 may use phase control to regulatethe output voltage provided that bi-directional voltage switches areused in at least the top side (for example Q5, Q6 of FIG. 71) or bottomside (for example Q7, Q8 of FIG. 71) of the rectifier circuit. In theconverters of FIGS. 71, 75, 79, 85, 89 and 93 bidirectional voltageswitches are provided by MOSFETs in series with blocking diodes (forexample Q5, Q6 and D1, D2 of FIG. 71) to block reverse voltage. If IGBTswitches are used then the blocking diodes could also be used. GTOswitches are themselves bidirectional and blocking diodes would not benecessary. Phase control may be used alone or in combination withswitching frequency control and/or phase shift control (whereapplicable).

[0337] It is noted that inductors L3, L4 are used to reduce theswitching loss of Q5, Q6 by filtering the waveform on the rectifiercircuit 7102 to a quasi-sinusoidal waveform. If the inductors L3, L4 arenot used then the circuit 7102 will operate; however, the switchinglosses for Q5, Q6 will be increased.

[0338] Referring to FIG. 72, a further alternate full-bridge parallelresonant converter 7201 is shown. It is different from the topology inFIG. 68B in that the resonant tank is changed from one inductor and onecapacitor to two capacitors (C1 and C2) and one inductor (L1). C1 is inparallel with the primary winding of the transformer T1 and in serieswith L1 and C2.

[0339] Other properties of this converter 7201 are similar to theconverter 6801 shown in FIG. 68B.

[0340] Referring to FIG. 73, a further alternate full-bridge parallelresonant converter 7301 is shown. It is different from the topologyshown in FIG. 69 in that the resonant tank is changed from one inductorand one capacitor to two capacitors (C1 and C2) and one inductor (L1).C1 is in parallel with the primary winding of the transformer T1 and inseries with L1 and C2.

[0341] Other properties of this converter 7301 are similar to theconverter 6901 shown in FIG. 69.

[0342] Referring to FIG. 74, a further alternate full-bridge parallelresonant converter 7401 is shown. It is different from the topologyshown in FIG. 70 in that the resonant tank is changed from one inductorand one capacitor to two capacitors (C1 and C2) and one inductor (L1).C1 is in parallel with the primary winding of the transformer T1 and inseries with L1 and C2.

[0343] Other properties of this converter 7401 are similar to theconverter 7001 shown in FIG. 70.

[0344] Referring to FIG. 75, a further alternate full-bridge parallelresonant converter 7501 is shown. It is different from the topologyshown in FIG. 71 in that the resonant tank is changed from one inductorand one capacitor to two capacitors (C1 and C2) and one inductor (L1).C1 is in parallel with the primary winding of the transformer T1 and inseries with L1 and C2.

[0345] Other properties of this converter 7501 are similar to theconverter 7101 shown in FIG. 71.

[0346] Referring to FIG. 76, a further alternate full-bridge parallelresonant converter 7601 is shown. It is different from the topology inFIG. 68B in that the resonant tank is changed from one inductor and onecapacitor to two capacitors (C1 and C2) and two inductors (L1 and L2).C1 and L2 are in parallel with the primary winding of the transformer T1and in series with L1 and C2.

[0347] Other properties of this converter 7601 are similar to theconverter 6801 shown in FIG. 68B.

[0348] Referring to FIG. 77, a further alternate full-bridge parallelresonant converter 7701 is shown. It is different from the topologyshown in FIG. 69 in that the resonant tank is changed from one inductorand one capacitor to two capacitors (C1 and C2) and two inductors (L1and L2). C1 and L2 are in parallel with the primary winding of thetransformer T1 and in series with L1 and C2.

[0349] Other properties of this converter 7701 are similar to theconverter 6901 shown in FIG. 69.

[0350] Referring to FIG. 78, a further alternate full-bridge parallelresonant converter 7801 is shown. It is different from the topologyshown in FIG. 70 in that the resonant tank is changed from one inductorand one capacitor to two capacitors (C1 and C2) and two inductors (L1and L2). C1 and L2 are in parallel with the primary winding of thetransformer T1 and in series with L1 and C2.

[0351] Other properties of this converter 7801 are similar to theconverter 7001 shown in FIG. 70.

[0352] Referring to FIG. 79, a further alternate full-bridge parallelresonant converter 7901 is shown. It is different from the topologyshown in FIG. 71 in that the resonant tank is changed from one inductorand one capacitor to two capacitors (C1 and C2) and two inductors (L1and L2). C1 and L2 are in parallel with the primary winding of thetransformer T1 and in series with L1 and C2.

[0353] Other properties of this converter 7901 are similar to theconverter 7101 shown in FIG. 71.

[0354] A number of non-isolated full-bridge series resonant converterembodiments of the converter 9601 will now be described. A non-isolatedfull-bridge series resonant converter 8001 is shown in FIG. 80. Theconverter 8001 has eight MOSFETs (Q1 to Q8), one transformer (T1), withone primary winding and one secondary winding, and an output capacitor(Cout). A resonant tank has one inductor (L1) and one capacitor (C1),the capacitor C1 is in series with the primary winding of T1 andinductor L1.

[0355] The converter 8001 has two circuits, a primary high side circuitand a secondary rectifier circuit. A high side circuit has a primarywinding of transformer T1, an auxiliary section (Q1 to Q4) and aresonant tank (L1, C1); while Q5 to Q8, the transformer T1 secondarywinding and Cout form a full-bridge rectifier circuit 8002.

[0356] Q1 to Q4 are primary switches. They can be implemented by, forexample. IRF7467 from International Rectifier. Q5 to Q8 are synchronousrectifier switches, they can be implemented by, for example, IRLR8103from International Rectifier. Q5 to Q8 can also be replaced by diodes.Other switches such as IGBT or GTO can also be used for Q1 to Q8.

[0357] Referring to FIG. 81, an alternate full-bridge series resonantconverter 8101 is shown. The converter 8101 has six MOSFETs (Q1 to Q6),one transformer (T1) with one primary winding and two secondarywindings, and one output capacitor (Cout). A resonant tank has oneinductor (L1) and one capacitor (C1), the capacitor C1 is in series withthe primary winding of T1 and inductor L1.

[0358] The converter 8101 has two circuits, a high side circuit and arectifier circuit 8101. High side circuit has a primary winding oftransformer T1, auxiliary section (Q1 to Q4) and resonant tank (L1, C1);while Q5 and Q6, the transformer T1 secondary winding and Cout form thecenter tapped transformer rectifier circuit.

[0359] Q1 to Q4 are primary switches. They can be implemented by, forexample, IRF7467 from International Rectifier. Q5, Q6 are synchronousrectifier switches, they can be implemented by, for example, IRLR8103from International Rectifier. Q5, Q6 can also be replaced by diodes.Other switches such as IGBT or GTO can also be used for Q1 to Q6.

[0360] A number of non-isolated half-bridge parallel resonant convertersembodiments of the converter 9601 will now be described. Referring toFIG. 82 a non-isolated half-bridge parallel resonant converter 8201 isshown. The converter 8201 has six MOSFETs (Q1 to Q6), one transformer(T1) with one primary winding and one secondary winding, two voltagedivider capacitors (C2 and C3), one output inductor (L2) and one outputcapacitor (Cout). A resonant tank has one inductor (L1) and onecapacitor (C1), the capacitor C1 is in parallel with the primary windingof T1 and in series with inductor L1.

[0361] The converter 8201 has two circuits, a high side circuit and arectifier circuit. High side circuit has the primary winding oftransformer T1, auxiliary section (Q1 to Q2 and two voltage dividercapacitors—C2 and C3) and a resonant tank (L1, C1); while Q3 to Q6,transformer T1 secondary winding, L2 and Cout form the full-bridgerectifier circuit 8202.

[0362] Q1 to Q2 are primary switches. They can be implemented by, forexample, IRF7467 from International Rectifier. Q3 to Q6 are synchronousrectifier switches, they can be implemented by, for example, IRLR8103from International Rectifier or be replaced by diodes. Other switchessuch as IGBT or GTO can also be used for Q1 to Q6.

[0363] Referring to FIG. 83, a further alternate half-bridge parallelresonant converter 8301 is shown. Its high side circuit is the same asthe topology shown in FIG. 82, but the rectifier circuit is changed fromfull-bridge rectifier in FIG. 82 to current doubler 8302. The currentdoubler 8302 has two MOSFETs (Q3 and Q4), two output inductors (L2 andL3), the secondary winding of the transformer T1 and an output capacitor(Cout).

[0364] Q3 and Q4 are synchronous rectifier switches, they can beimplemented by, for example, IRLR8103 from International Rectifier or bereplaced by diodes. Other switches such as IGBT or GTO can also be usedfor Q1 to Q4.

[0365] Referring to FIG. 84, a further alternate half-bridge parallelresonant converter 8401 is shown. Its high side circuit is the same asthe topology shown in FIG. 82, but the rectifier circuit is changedfroma full-bridge rectifier in FIG. 82 to a center tapped transformerrectifier 8402. The center tapped transformer rectifier 8402 has twoMOSFETs (Q3 and Q4), one output inductor (L2), two secondary windings ofthe transformer T1 and an output capacitor (Cout).

[0366] Q3 and Q4 are synchronous rectifier switches, they can beimplemented by, for example, IRLR8103 from International Rectifier or bereplaced by diodes. Other switches such as IGBT or GTO can also be usedfor Q1 to Q4.

[0367] Referring to FIG. 85, a half-bridge parallel resonant converter8501 is shown. Its high side circuit is the same as the topology shownin FIG. 82, but the rectifier circuit is changed from a full-bridgerectifier in FIG. 82 to a phase control rectifier 8502. The phasecontrol rectifier 8502 has four MOSFETs (Q3 to Q6), three diodes (D1 toD3), three inductors (L2 L3 and L4), one secondary winding of thetransformer T1 and an output capacitor (Cout).

[0368] D1 to D3 can also be replaced with MOSFETs to reduce conductionloss. Q3 to Q6 are synchronous rectifier switches, they can beimplemented by, for example, IRLR8103 from International Rectifier.Other switches such as IGBT or GTO can also be used for Q1 to Q6.

[0369] Referring to FIG. 86, a further alternate half-bridge parallelresonant converter 8601 is shown. It is different from the topologyshown in FIG. 82 in that the resonant tank is changed from one inductorone capacitor to two capacitors (C1 and C4) and one inductor L1. C1 isin parallel with the primary winding of the transformer T1 and in serieswith L1 and C4.

[0370] Other properties of this topology are similar to the converter8201 in FIG. 82.

[0371] Referring to FIG. 87, a half-bridge parallel resonant converter8701 is shown. It is different from the topology shown in FIG. 83 inthat the resonant tank is changed from one inductor and one capacitor totwo capacitors (C1 and C4) and one inductor L1. C1 is in parallel withthe primary winding of the transformer T1 and in series with L1 and C4.

[0372] Other properties of this topology are similar to the topology inFIG. 83.

[0373] Referring to FIG. 88, a further alternate half-bridge parallelresonant converter 8801 is shown. It is different from the topologyshown in FIG. 84 in that the resonant tank is changed from one inductorand one capacitor to two capacitors (C1 and C4) and one inductor L1. C1is in parallel with the primary winding of the transformer T1 and inseries with L1 and C4.

[0374] Other properties of this topology are similar to the topology inFIG. 84.

[0375] Referring to FIG. 89, a further alternate half-bridge parallelresonant converter 8901 is shown. It is different from the topologyshown in FIG. 85 in that the resonant tank is changed from one inductorand one capacitor to two capacitors (C1 and C4) and one inductor L1. C1is in parallel with the primary winding of the transformer T1 and inseries with L1 and C4.

[0376] Other properties of this topology are similar to the topology inFIG. 85.

[0377] Referring to FIG. 90, a half-bridge parallel resonant converter9001 is shown. It is different from the topology shown in FIG. 82 inthat the resonant tank is changed from one inductor and one capacitor totwo capacitors (C1 and C4) and two inductors (L1 and L2). C1 and L2 arein parallel with the primary winding of the transformer T1 and in serieswith L1 and C4.

[0378] Other properties of this topology are similar to the topology inFIG. 82.

[0379] Referring to FIG. 91, a further alternate half-bridge parallelresonant converter 9101 is shown. It is different from the topologyshown in FIG. 83 in that the resonant tank is changed from one inductorand one capacitor to two capacitors (C1 and C4) and two inductors (L1and L2). C1 and L2 are in parallel with the primary winding of thetransformer T1 and in series with L1 and C4.

[0380] Other properties of this topology are similar to the topology inFIG. 83.

[0381] Referring to FIG. 92, a half-bridge parallel resonant converter9201 is shown. It is different from the topology in FIG. 84 in that theresonant tank is changed from one inductor and one capacitor to twocapacitors (C1 and C4) and two inductors (L1 and L2). C1 and L2 are inparallel with the primary winding of the transformer T1 and in serieswith L1 and C4.

[0382] Other properties of this topology are similar to the topology inFIG. 84.

[0383] Referring to FIG. 93, a further alternate half-bridge parallelresonant converter 9301 is shown. It is different from the topology inFIG. 85 in that the resonant tank is changed from one inductor and onecapacitor to two capacitors (C1 and C4) and two inductors (L1 and L2).C1 and L2 are in parallel with the primary winding of the transformer T1and in series with L1 and C4.

[0384] Other properties of this topology are similar to the topology inFIG. 85.

[0385] A number of non-isolated half-bridge series resonant converterembodiments of the converter 9601 will now be described. Referring toFIG. 94, a non-isolated half-bridge series resonant converter 9401 isshown. The converter 9401 has six MOSFETs (Q1 to Q6), one transformer(T1) with one primary winding and one secondary winding, two voltagedivider capacitors (C2 and C3), and one output capacitor (Cout). Aresonant tank has one inductor L1 and one capacitor C1, the capacitor C1is in series with the primary winding of T1 and inductor L1.

[0386] The converter 9401 has two circuits, a high side circuit and arectifier circuit. The high side circuit has the primary winding oftransformer T1, an auxiliary section (Q1 and Q2 and two voltage dividercapacitors C2 and C3) and a resonant tank (L1, C1); while Q3 to Q6,transformer T1 secondary winding and Cout form the full-bridge rectifiercircuit 9402.

[0387] Q1 and Q2 are primary switches. They can be implemented by, forexample, IRF7467 from International Rectifier. Q3 to Q6 are synchronousrectifier switches, they can be implemented by, for example, IRLR8103from International Rectifier. Q3 to Q6 can also be replaced by diodes.Other switches such as IGBT or GTO can also be used for Q1 to Q6.

[0388] For this topology the duty cycle of Q5 and Q6 can be controlledto change the output voltage.

[0389] Referring to FIG. 95, an alternate half-bridge series resonantconverter 9501 is shown. The converter 9501 has four MOSFETs (Q1 to Q4),one transformer (T1) with one primary winding and two secondarywindings, two voltage divider capacitors (C2 and C3), and one outputcapacitor (Cout). A resonant tank has one inductor (L1) and onecapacitor (C1), the capacitor C1 is in series with the primary windingof T1 and inductor L1.

[0390] The converter 9501 has two circuits, a high side circuit and arectifier circuit. The high side circuit has the primary winding oftransformer T1, an auxiliary section (Q1 to Q2, two voltage dividercapacitors C2 and C3) and a resonant tank (L1, C1); while Q3 and Q4,transformer T1 secondary winding and Cout form the center tappedtransformer rectifier circuit.

[0391] Q1 Q2 are primary switches. They can be implemented by, forexample, IRF7467 from International Rectifier. Q3 Q4 are synchronousrectifier switches, they can be implemented by, for example, IRLR8103from International Rectifier. Q3,Q4 can also be replaced by diodes.Other switches such as IGBT or GTO can also be used for Q1 to Q4.

[0392] For this topology the duty cycle of Q3 and Q4 can be controlledto change the output voltage.

[0393] Using similar techniques other resonant converters can also bebuilt. This is shown in the block diagram in FIG. 68A. By usingdifferent and/or multiple high side circuits, different resonant tanks,multiple primary and/or secondary windings, and/or different and/ormultiple rectifier circuits, different converters can be formed. Thosecircuits and tanks include such high side circuits, resonant tanks andrectifier circuits as would be understood by those skilled in the art tobe applicable for these purposes, including those circuits described assuch anywhere in this description. The techniques described herein forcreating non-resonant tank converters using multiple high side circuitsand multiple rectifier circuits are equally applicable to resonant tankconverters.

[0394] The advantages of these topologies include, because the primaryand secondary sides (high side and rectifier circuits) are not isolated,the primary and secondary sides being coupled directly, resulting in anincrease in efficiency or a decrease in power loss during conversion.This can significantly improve the life of the converter, as thejunction temperature of semiconductors is reduced. In addition, this canmake the converter size smaller. It can also simplify the mechanicaldesign of the computer motherboard, which can reduce the cost.

[0395] Another advantage of these converters is that they can operatewith very high switching frequency because the switches can operate inZVS or ZCS (“zero current switching”) mode, with a resulting reductionin switch losses.

[0396] As previously mentioned, the control methods for use withresonant tank converters can include switching frequency control, phaseshift control, and phase control, alone or together.

[0397] It will be understood by those skilled in the art that thisdescription is made with reference to the preferred embodiment and thatit is possible to make other embodiments employing the principles of theinvention which fall within its spirit and scope as defined by thefollowing claims.

We claim:
 1. A DC-DC converter for use with a DC power source having aDC voltage across a first voltage source output and a second voltagesource output and with a load, the converter comprising: a. an input foraccepting the DC voltage, the input having a first voltage input and asecond voltage input, b. an output for outputting a converted DCvoltage, the output having a first voltage output and a second voltageoutput, c. a high side circuit including a first primary winding of afirst transformer and an auxiliary section, the high side circuitconnected between the first voltage input and the first voltage output,d. a rectifier circuit having a first secondary winding of the firsttransformer, the rectifier circuit connected between the first voltageoutput and the second voltage output, and e. an output capacitorconnected between the first voltage output and the second voltage outputand across the rectifier circuit, wherein an output converter DC voltagebetween the first voltage output and the second voltage output has thesame polarity as a DC voltage input between the first voltage input andthe second voltage input, wherein the auxiliary section is for causingthe first transformer to transfer power from the first primary windingto the first secondary winding and to operate without saturation,wherein the high side circuit has a high side circuit output connectedsuch that current flowing through the first primary winding is directedbetween the high side circuit output and the first voltage output,wherein the rectifier circuit is for converting output of the firstsecondary winding into a one-direction waveform and converting theone-direction waveform into a DC voltage, and wherein the outputcapacitor is for filtering the converted DC voltage.
 2. The converter ofclaim 1, wherein the auxiliary section comprises switches for repeatedlyconnecting and disconnecting the primary winding from the input, andallows for resetting of the first transformer.
 3. The converter of claim2, wherein the auxiliary section comprises a combination of switches andcapacitors.
 4. The converter of claim 1, wherein the auxiliary sectioncomprises four switches.
 5. The converter of claim 4, wherein eachswitch is a MOSFET.
 6. The converter of claim 1, wherein the auxiliarysection comprises a first switch connected between a first side of thefirst primary winding and the first voltage input, a second switchconnected between a second side of the first primary winding and thefirst voltage input, a third switch connected between the first side ofthe first primary winding and the high side circuit output, and a fourthswitch connected between the second side of the first primary windingand the high side circuit output.
 7. The converter of claim 6, whereineach switch has an input for a gate drive signal for controlling theoperation of the switch.
 8. The converter of claim 7, wherein the gatedrive signals repeatedly turn on and turn off the first and fourthswitch, as well as turn on and off the second and third switch.
 9. Theconverter of claim 1, wherein the auxiliary section comprises a firstswitch connected between a first side of the first primary winding andthe first voltage input, a first capacitor connected between a secondside of the first primary winding and the first voltage input, a secondswitch connected between the first side of the first primary winding andthe high side circuit output, and a second capacitor connected betweenthe second side of the first primary winding and the high side circuitoutput.
 10. The converter of claim 9, wherein each switch has an inputfor a gate drive signal for controlling the operation of the switch. 11.The converter of claim 10, further comprising gate drive signals adaptedto repeatedly turn on and turn off the first switch and the secondswitch, whereby the first transformer can be reset from the capacitors.12. The converter of claim 10, wherein the capacitors are large enoughthat the voltage across the capacitors will not change significantlyduring normal operation of the converter.
 13. The converter of claim 1,wherein the auxiliary section comprises a first switch connected betweena first side of the first primary winding and the first voltage input, afirst diode connected between a second side of the first primary windingand the first voltage input for forward conduction from the second sideof the first primary winding to the first voltage input, a second switchconnected between the second side of the first primary winding and thehigh side circuit output, and a second diode connected between the firstside of the first primary winding and the high side circuit output forforward conduction from the high side circuit output to the first sideof the first primary winding.
 14. The converter of claim 13, whereineach switch has an input for a gate drive signal for controlling theoperation of the switch.
 15. The converter of claim 14, furthercomprising gate drive signals adapted to repeatedly turn on and turn offthe first switch and the second switch, whereby the first transformercan be reset by current flowing through the first and second diodes. 16.The converter of claim 1, wherein the auxiliary section comprises afirst side of the first primary winding connected to the first voltageinput, a first switch connected between the first side of the firstprimary winding and the first side of the first capacitor, the secondside of the first capacitor connected to the second side of the firstprimary winding, a second switch connected between the second side ofthe first primary winding and the high side circuit output.
 17. Theconverter of claim 16, wherein each switch has an input for a gate drivesignal for controlling the operation of the switch.
 18. The converter ofclaim 17, further comprising gate drive signals adapted to repeatedlyturn on the first switch, while turning off the second switch, and turnoff the first switch, while turning on the second switch, whereby thefirst transformer can be reset from the first capacitor.
 19. Theconverter of claim 1, wherein the rectifier circuit further comprises acombination of inductors and switches, wherein the switches are forconverting alternating voltage in the first secondary winding intopulsating one-direction voltage and the inductors are for convertingpulsating one-direction voltage into DC voltage.
 20. The converter ofclaim 1, wherein the rectifier circuit further comprises a combinationof inductors and diodes, wherein the diodes are for converting pulsatingalternating voltage in the first secondary winding into pulsatingone-direction voltage and the inductors are for converting pulsatingone-direction voltage into DC voltage.
 21. The converter of claim 1,wherein the rectifier circuit further comprises a first rectifier switchconnected between the second voltage output and a first side of thefirst secondary winding, a second rectifier switch connected between asecond side of the first secondary winding and the second voltageoutput, a first inductor connected between the first side of the firstsecondary winding and the first voltage output, and a second inductorconnected between the second side of the first secondary winding and thefirst voltage output.
 22. The converter of claim 21, wherein each switchhas an input for a gate drive signal for controlling the operation ofthe switch.
 23. The converter of claim 22, further comprising gate drivesignals adapted to switch the first and second rectifier switches toconvert bi-directional AC voltage at the first secondary winding intoone-direction pulsating voltage.
 24. The converter of claim 1, whereinthe rectifier circuit further comprises a first rectifier switchconnected between the second voltage output and a first side of thefirst secondary winding, a second rectifier switch connected between asecond side of the first secondary winding and the second voltageoutput, and a first inductor connected between the first side of thefirst secondary winding and the first voltage output not in series withthe second rectifier switch.
 25. The converter of claim 1, wherein therectifier circuit further comprises first and second rectifier diodesand a first inductor, wherein the first diode is connected between afirst side of the first secondary winding and the first inductor, andthe inductor is further connected between the first diode and the firstvoltage output, for forward conduction from the secondary windingthrough the inductor, and the second diode is connected between (a) apoint between the second side of the first secondary winding and thesecond voltage output and (b) a point between the first inductor andfirst diode, also for forward conduction from the secondary windingthrough the inductor.
 26. The converter of claim 1, wherein therectifier circuit further comprises a second secondary winding, firstand second rectifier diodes and a first inductor, wherein a second sideof the first secondary winding is connected to a first side of thesecond secondary winding and the second voltage output, and the firstdiode is connected between a first side of the first secondary windingand the first inductor, and the inductor is further connected betweenthe first diode and the first voltage output, for forward conductionfrom the secondary winding through the inductor, and the second diode isconnected between the second side of the second secondary winding and apoint between the first inductor and first diode, also for forwardconduction from the secondary winding through the inductor.
 27. Theconverter of claim 1, wherein the rectifier circuit further comprises asecond secondary winding, first and second rectifier switches and afirst inductor, wherein a second side of the first secondary winding isconnected to a first side of the second secondary winding and theinductor which is further connected to the first voltage output, and thefirst rectifier switch is connected between a first side of the firstsecondary winding and the second voltage output, and the secondrectifier switch is connected between a second side of the secondsecondary winding and the second voltage output.
 28. The converter ofclaim 21, wherein the first and second rectifier switches, the firstsecondary winding and the first and second inductors are comprisedwithin a first rectifier section, and the rectifier circuit furthercomprises a second rectifier section similar to the first rectifiersection, and the first and second rectifier sections are connected inparallel with one another and with the output capacitor and the output.29. The converter of claim 26, wherein the first and second diodes, thefirst secondary and second secondary windings and the inductor arecomprised within a first rectifier section, and the rectifier circuitfurther comprises a second rectifier section similar to the firstrectifier section, and the first and second rectifier sections areconnected in parallel with one another and with the output capacitor andthe output.
 30. The converter of claim 27, wherein the first and secondrectifier switches, the first secondary and second secondary windingsand the inductor are comprised within a first rectifier section, and therectifier circuit further comprises a second rectifier section similarto the first rectifier section, and the first and second rectifiersections are connected in parallel with one another and with the outputcapacitor and the output.
 31. The converter of claim 1, furthercomprising a second converter similar to the converter of claim 1,wherein the two converters are connected in parallel with one another attheir respective inputs and outputs.
 32. The converter of claim 31,wherein the output capacitors of the two converters are combined as asingle physical capacitor.
 33. The converter of claim 31, wherein thetwo converters have inputs for interleaved gate drive signals, wherebyvoltage ripple incoming to the output capacitor is reduced, allowing forreduction in the size of the output capacitor.
 34. The converter ofclaim 1, further comprising a second transformer and wherein the highside circuit further comprises a second primary winding of the secondtransformer, and first and second second primary switches, wherein thefirst second primary switch is connected between the first voltage inputand a first side of the second primary winding, and the second secondprimary switch is connected between the first side of the second primarywinding and the first voltage output, and a second side of the secondprimary winding is connected to a side of the first primary winding, andwherein the rectifier circuit comprises a second rectifier circuitsimilar to and connected in parallel with the first rectifier circuit,wherein the second rectifier circuit includes a second second rectifiersecondary of the second transformer.
 35. The converter of claim 34,wherein the second primary switches have inputs for gate drive signalsfor operating the second primary winding out of phase with the firstprimary winding.
 36. The converter of claim 1, further comprising asecond transformer and wherein the high side circuit further comprises asecond primary winding of the second transformer, and first and secondsecond primary switches, wherein the first second primary switch isconnected between the first voltage input and a first side of the secondprimary winding, and the second second primary switch is connectedbetween the first side of the second primary winding and the high sidecircuit output, and a second side of the second primary winding isconnected to a side of the first primary winding, and wherein therectifier circuit comprises a second rectifier secondary winding, asecond rectifier inductor and a second rectifier switch, wherein a firstside of the second rectifier secondary winding is connected to a firstside of the first secondary winding and the second rectifier switch isconnected between a second side of the second secondary winding and thesecond voltage output, and the inductor is connected between the secondside of the second secondary winding and the high side circuit output,not in series with the second rectifier switch.
 37. The converter ofclaim 36, wherein the second primary switches have inputs for gate drivesignals for operating the second primary winding out of phase with thefirst primary winding, and the first and second rectifier circuits haveinputs for gate drive signals for operating the rectifier circuitsecondary windings phase shifted from one another.
 38. The converter ofclaim 1, further comprising a second transformer and a thirdtransformer, wherein the high side circuit further comprises a secondprimary winding of the second transformer, and first and second secondprimary switches, and wherein the first second primary switch isconnected between the first voltage input and a first side of the secondprimary winding, and the second second primary switch is connectedbetween the first side of the second primary winding and the high sidecircuit output, and a second side of the second primary winding isconnected to a side of the first primary winding, and wherein therectifier circuit comprises a second rectifier circuit and a thirdrectifier circuit each similar to and connected in parallel with thefirst rectifier circuit, wherein the second rectifier circuit includes asecond second rectifier secondary of the second transformer and thethird rectifier circuit includes a third third rectifier secondary ofthe third transformer.
 39. The converter of claim 38, wherein the secondprimary switches have inputs for gate drive signals for operating thesecond primary winding out of phase with the first primary winding, andno additional drive components are added for the third primary winding,wherein the converter has gate drive inputs for operating the thirdprimary winding partially in phase with the first primary winding andpartially in phase with the second primary winding.
 40. The converter ofclaim 1, wherein the first voltage input is for accepting a DC potentialthat is positive when compared to a DC potential for acceptance by thesecond voltage input.
 41. The converter of claim 40, wherein the DCpotential of the first voltage output is positive when compared to theDC potential of the second voltage output.
 42. The converter of claim 1,wherein the input voltage is within a range of 10.8 volts DC to 13.2volts DC, and the out put voltage is within a range of 0.8 volts DC to1.6 volts DC.
 43. The converter of claim 1, wherein the duty cycle isbetween 40% and 60%.
 44. The converter of claim 1, wherein the dutycycle is approximately 50%.
 45. The converter of claim 31, wherein theduty cycle of each of the first and second converters is between 40% and60%.
 46. The converter of claim 31, wherein the duty cycle of each ofthe first and second converters is approximately 50%.
 47. The converterof claim 38, wherein the duty cycle of each of the first, second andthird transformers is approximately 33⅓%.
 48. The converter of claim 1,further comprising a second high side circuit similar to the first highside circuit, connected in parallel with the first high side circuit,and a second rectifier circuit connected in parallel with the firstrectifier circuit.
 49. The converter of claim 48, further comprisinginputs for drive signals to operate the first high side circuit and thefirst rectifier circuit out of phase with the second high side circuitand the second rectifier circuit, respectively.
 50. The converter ofclaim 48, further comprising gate drive signals for operating the firsthigh side circuit and the first rectifier circuit out of phase with thesecond high side circuit and the second rectifier circuit, respectively.51. The converter of claim 1, further comprising a current sensor inseries with the high side circuit.
 52. The converter of claim 51,wherein current sensed at the current sensor is for use in determiningthe timing of gate drive signals for operating the high side circuit.53. The converter of claim 1, wherein the output of the first secondarywinding is a pulsating voltage and the one-direction waveform is aone-direction voltage.
 54. The converter of claim 1, wherein the highside circuit further comprises a resonant tank.
 55. The converter ofclaim 54, wherein the resonant tank comprises a first capacitor inparallel with the first primary winding and a first inductor in serieswith the first primary winding between the first primary winding and theauxiliary circuit.
 56. The converter of claim 55, wherein the resonanttank further comprises a second capacitor in series with the firstinductor between the first primary circuit and the auxiliary circuit.57. The converter of claim 56, wherein the resonant tank furthercomprises a second inductor in parallel with the first primary windingand the first capacitor.
 58. The converter of claim 54, wherein theresonant tank comprises a first inductor and a first capacitor in serieswith one another between the first primary winding and the auxiliarycircuit.
 59. The converter of claim 54, wherein the rectifier circuitcomprises a full-bridge rectifier.
 60. The converter of claim 54,wherein the rectifier circuit comprises a half-bridge rectifier.
 61. Theconverter of claim 54, comprising switches that are controlled byswitching frequency control to regulate the output voltage.
 62. Theconverter of claim 54, wherein the auxiliary section is a full-bridgeauxiliary section.
 63. The converter of claim 62, wherein switcheswithin the auxiliary section are controlled by phase shift control. 64.The converter of claim 59, wherein switches within the rectifier circuitare controlled by phase control.
 65. A method of operating a DC-DCconverter for use with a DC power source having a DC voltage across afirst voltage source output and a second voltage source output and witha load, the converter comprising: a. an input for accepting the DCvoltage, the input having a first voltage input and a second voltageinput, b. an output for outputting a converter DC voltage, the outputhaving a first voltage output and a second voltage output, c. a highside circuit including a first primary winding of a first transformerand an auxiliary section, the high side circuit connected between thefirst voltage input and the second voltage output, d. a rectifiercircuit having a first secondary winding of the first transformer, therectifier circuit connected between the first voltage output and thesecond voltage output, and e. an output capacitor connected between thefirst voltage output and the second voltage output and across therectifier circuit, wherein an output converter DC voltage between thefirst voltage output and the second voltage output has the same polarityas a DC voltage input between the first voltage input and the secondvoltage input, wherein the auxiliary section is for causing the firsttransformer to transfer power from the first primary winding to thefirst secondary winding and to operate without saturation, wherein thehigh side circuit has a high side circuit output connected such thatcurrent flowing through the first primary winding is directed betweenthe high side circuit output and the first voltage output, wherein therectifier circuit is for converting output of the first secondarywinding into a one-direction waveform and converting the one-directionwaveform into a DC voltage, and wherein the output capacitor is forfiltering the converted DC voltage, the method comprising the steps of:driving the auxiliary section to cause the first transformer to transferpower from the first primary winding to the first secondary winding,while at the same time driving the auxiliary section to cause thetransformer to operate without saturation.